Method and system for fast acquisition of ultra-wideband signals

ABSTRACT

A system and method are provided that can detect any part of a multipath impulse radio signal. More specifically, the method compares a template pulse train and the multipath impulse radio signal to obtain a comparison result. The system performs a threshold check on the comparison result. If the comparison result passes the threshold check, the system locks onto any part of the multipath impulse radio signal including a direct path part and at least one multipath reflection part. The system may also perform a quick check, a synchronization check, and/or a command check of the multipath impulse radio signal.

CROSS REFERENCE TO COPENDING APPLICATIONS

This application is a Continuation of U.S. application Ser. No.11/195,402 that was filed on Aug. 2, 2005, which is a Continuation ofU.S. application Ser. No. 10/356,995 filed Feb. 3, 2003 (issued as U.S.Pat. No. 6,925,109), which is a Continuation-In-Part to U.S. applicationSer. No. 09/538,292 that was filed on Mar. 29, 2000 (issued as U.S. Pat.No. 6,556,621), all of which are hereby incorporated by referenceherein.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates in general to the communications field and, moreparticularly, to a system and method capable of locking ontoultra-wideband signals in a multipath environment.

2. Description of Related Art

Recent advances in communications technology have enabled an emergingand revolutionary ultra-wideband technology (UWB) called impulse radiocommunications systems (hereinafter called impulse radio). Basic impulseradio transmitters emit short pulses approaching a Gaussian monocyclewith tightly controlled pulse-to-pulse intervals. Impulse radio systemstypically use pulse position modulation, which is a form of timemodulation where the value of each instantaneous sample of a modulatingsignal is caused to modulate the position of a pulse in time.

For impulse radio communications, the pulse-to-pulse interval is variedon a pulse-by-pulse basis by two components: an information componentand a pseudo-random code component. Unlike direct sequence spreadspectrum systems, the pseudo-random code for impulse radiocommunications is not necessary for energy spreading because themonocycle pulses themselves have an inherently wide bandwidth. Instead,the pseudo-random code of an impulse radio system is used forchannelization, energy smoothing in the frequency domain, andinterference suppression.

Generally speaking, an impulse radio receiver is a direct conversionreceiver with a cross-correlator front end. The front end coherentlyconverts an electromagnetic pulse train of monocycle pulses to abaseband signal in a single stage. The data rate of the impulse radiotransmission is typically a fraction of the periodic timing signal usedas a time base. Because each data bit modulates the time position ofmany pulses of the periodic timing signal, this yields a modulated,coded timing signal that comprises a train of identically shaped pulsesfor each single data bit. As an option, the impulse radio receiver canintegrate multiple pulses to recover the transmitted information.

In a multi-user environment, impulse radio depends, in part, onprocessing gain to achieve rejection of unwanted signals. Because of theextremely high processing gain achievable with impulse radio, muchhigher dynamic ranges are possible than are commonly achieved with otherspread spectrum methods.

When receiving an ultra-wideband signal as part of a one-waycommunication system (e.g., a pager) or a two-way communication system(e.g., a wireless telephone), a problem exists as to how to detect thetransmitted impulse radio signal, and more particularly, how to acquireand lock onto the impulse radio signal being transmitted. This problemexists independent of how the received ultra-wideband signal is used,such as in a one-way or two-way communication system.

Previous approaches to solving this problem are discussed in thefollowing commonly owned patents, which are incorporated by reference:U.S. Pat. No. 5,832,035, issued Nov. 3, 1998 to Fullerton, and U.S. Pat.No. 5,677,927, issued Oct. 14, 1997. The present invention presentsanother approach to solving this problem.

BRIEF DESCRIPTION OF THE INVENTION

An aspect of the invention is to provide a system and method fordetecting an impulse radio signal.

Another aspect of the invention is to provide a system and method forlocking onto any part of a received multipath impulse radio signalincluding one or more of a direct path part and at least one multipathreflection part.

The above aspects and advantages of the present invention are achievedby a method and system for fast lock and acquisition of ultra-widebandsignals.

In one embodiment of the present invention, a system and method areprovided that can detect any part of a multipath impulse radio signal.More specifically, the method compares a template pulse train and themultipath impulse radio signal to obtain a comparison result. The systemperforms a threshold check on the comparison result. If the comparisonresult passes the threshold check, the system locks onto any part of themultipath impulse radio signal including a direct path part and at leastone multipath reflection part. The system may also perform a quickcheck, a synchronization check, and/or a command check of the multipathimpulse radio signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is described with reference to the accompanyingdrawings. In the drawings, like reference numbers indicate identical orfunctionally similar elements.

FIG. 1A illustrates a representative Gaussian Monocycle waveform in thetime domain, which is the first derivative of a Gaussian pulse;

FIG. 1B illustrates the frequency domain amplitude of the GaussianMonocycle of FIG. 1A;

FIG. 1C represents the second derivative of the Gaussian pulse;

FIG. 1D represents the third derivative of the Gaussian pulse;

FIG. 1E represents the Correlator Output vs. the Relative Delay of ameasured pulse signal;

FIG. 1F depicts the frequency domain amplitude of the Gaussian family ofthe Gaussian Pulse and the first, second, and third derivative;

FIG. 2A illustrates a pulse train comprising pulses as in FIG. 1A;

FIG. 2B illustrates the frequency domain amplitude of the waveform ofFIG. 2A;

FIG. 2C illustrates the pulse train spectrum;

FIG. 2D is a plot of the Frequency vs. Energy;

FIG. 3 illustrates the cross-correlation of two codes graphically asCoincidences vs. Time Offset;

FIGS. 4A-4E illustrate five modulation techniques to include: Early-LateModulation; One of Many Modulation; Flip Modulation; Quad FlipModulation; and Vector Modulation;

FIG. 5A illustrates representative signals of an interfering signal, acoded received pulse train and a coded reference pulse train;

FIG. 5B depicts a typical geometrical configuration giving rise tomultipath received signals;

FIG. 5C illustrates exemplary multipath signals in the time domain;

FIG. 5D represents a signal plot of an idealized UWB received pulse withno multipath;

FIG. 5E represents a signal plot of an idealized UWB received pulse inmoderate multipath;

FIG. 5F represents a signal plot of an idealized UWB received pulse insevere multipath;

FIG. 5G illustrates the Rayleigh fading curve associated withnon-impulse radio transmissions in a multipath environment;

FIG. 5H illustrates a plurality of multipaths with a plurality ofreflectors from a transmitter to a receiver;

FIG. 5I graphically represents signal strength as volts vs. time in adirect path and multipath environment;

FIG. 6 illustrates a representative impulse radio transmitter functionaldiagram;

FIG. 7 illustrates a representative impulse radio receiver functionaldiagram;

FIG. 8A illustrates a representative received pulse signal at the inputto the correlator;

FIG. 8B illustrates a sequence of representative impulse signals in thecorrelation process;

FIG. 8C illustrates the output of the correlator for each of the timeoffsets of FIG. 8B;

FIG. 9 illustrates a flow diagram for fast lock and acquisition of animpulse radio signal using the invention;

FIG. 10A illustrates two exemplary forms of a received impulse radiosignal;

FIG. 10B illustrates a template pulse train;

FIG. 10C illustrates shifting a template pulse train by an offset;

FIG. 10D is an alternative presentation of the information in FIG. 10Cusing a cyclic representation;

FIG. 11A illustrates an exemplary template pulse train that can have afixed frame time or a non-fixed time frame;

FIG. 11B illustrates an exemplary template pulse train that has a fixedframe time;

FIG. 11C illustrates a received impulse radio signal compared to atemplate pulse train that was shifted in accordance with an exemplaryscanning process;

FIG. 11D is a simplified cyclic diagram showing the relative samplingpositions in a fine step scanning process;

FIG. 11E is a simplified cyclic diagram showing the relative samplingpositions in a coarse step scanning process;

FIG. 11F illustrates a received impulse radio signal compared to atemplate pulse train that was shifted in accordance with a reversebinary scanning process;

FIG. 11G illustrates in greater detail the reverse binary scanningprocess of FIG. 11F;

FIG. 11H illustrates delays between scans of a reverse binary scanningprocess in accordance with one embodiment of the invention;

FIG. 11I illustrates an optimized reverse binary scanning order inaccordance with one embodiment of the invention;

FIG. 11J illustrates a timing diagram for two receiving blockscollaboratively performing a reverse binary search process;

FIG. 11K illustrates a timing diagram for two receiving blockscollaborative performing a reverse binary search process using anoptimized reverse binary scanning order in accordance with oneembodiment of the invention;

FIG. 11L illustrates a timing diagram for a reverse binary searchprocess involving scanning across received pulse train boundaries;

FIG. 11M illustrates a block diagram of a modification to FIG. 7 inaccordance with one embodiment of the invention;

FIG. 11N illustrates a switch-based inverting pulse summation apparatusin accordance with one embodiment of the invention;

FIG. 11O illustrates a mixer-based inverting pulse summation apparatusin accordance with one embodiment of the invention;

FIG. 11P illustrates step size selection in a multipath environment;

FIG. 12A illustrates a summation of multiple pulse signals leading to adigital bit decision;

FIG. 12B illustrates a statistical probability density relating to theoutput and final value of the summation of FIG. 12A;

FIG. 13A illustrates a flow diagram for a discrete system tracking loopto maintain signal timing;

FIG. 13B illustrates a flow diagram for a continuous system trackingloop to maintain signal timing;

FIG. 13C illustrates a flow diagram for an alternative continuous systemtracking loop to maintain signal timing;

FIG. 14 illustrates a flow diagram for the quick check;

FIG. 15A illustrates acquisition data;

FIG. 15B illustrates a flow diagram for the synchronization check;

FIG. 16 illustrates a flow diagram for the command check;

FIG. 17 illustrates a block diagram for a system capable of implementingthe invention; and

FIG. 18 illustrates a receiver utilizing two correlators to provideseparate signals for tracking and data.

DETAILED DESCRIPTION OF THE INVENTION

Overview of the Invention

The present invention will now be described more fully in detail withreference to the accompanying drawings, in which the preferredembodiments of the invention are shown. This invention should not,however, be construed as limited to the embodiments set forth herein;rather, they are provided so that this disclosure will be thorough andcomplete and will fully convey the scope of the invention to thoseskilled in art. Like numbers refer to like elements throughout.

Ultra Wideband Technology Overview

Ultra Wideband is an emerging RF technology with significant benefits incommunications, radar, positioning and sensing applications. Earlierthis year (2002), the Federal Communications Commission (FCC) recognizedthese potential benefits to the consumer and issued the first rulemakingenabling the commercial sale and use of products based on Ultra Widebandtechnology in the United States of America. The FCC adopted a definitionof Ultra Wideband to be a signal that occupies a fractional bandwidth ofat least 0.25, or 0.5 GHz bandwidth at any center frequency. The 0.25fractional bandwidth is more precisely defined as:

${{FBW} = \frac{2\left( {f_{h} - f_{l}} \right)}{f_{h} + f_{1}}},$where FBW is the fractional bandwidth, f_(h) is the upper band edge andf_(l) is the lower band edge, the band edges being defined as the 10 dBdown point in spectral density.

There are many approaches to UWB including impulse radio, directsequence CDMA, ultra wideband noise radio, direct modulation of ultrahigh-speed data, and other methods. The present invention has its originin ultra wideband impulse radio and will have significant applicationthere, but it has potential benefit and application beyond impulse radioto other forms of ultra wideband and beyond ultra wideband toconventional radio systems as well. Nonetheless, it is useful todescribe the invention in relation to impulse radio to understand thebasics and then expand the description to the extensions of thetechnology.

The following is an overview of impulse radio as an aid in understandingthe benefits of the present invention.

Impulse radio has been described in a series of patents, including U.S.Pat. Nos. 4,641,317 (issued Feb. 3, 1987), 4,813,057 (issued Mar. 14,1989), 4,979,186 (issued Dec. 18, 1990), and 5,363,108 (issued Nov. 8,1994) to Larry W. Fullerton. A second generation of impulse radiopatents includes U.S. Pat. Nos. 5,677,927 (issued Oct. 14, 1997),5,687,169 (issued Nov. 11, 1997), 5,764,696 (issued Jun. 9, 1998),5,832,035 (issued Nov. 3, 1998), and 5,969,663 (issued Oct. 19, 1999) toFullerton et al, and 5,812,081 (issued Sep. 22, 1998), and 5,952,956(issued Sep. 14, 1999) to Fullerton, which are incorporated herein byreference.

Uses of impulse radio systems are described in U.S. Pat. No. 6,177,903(issued Jan. 23, 2001) titled, “System and Method for IntrusionDetection using a Time Domain Radar Array” and U.S. Pat. No. 6,218,979(issued Apr. 17, 2001) titled “Wide Area Time Domain Radar Array”, bothof which are assigned to the assignee of the present invention, and areincorporated herein by reference.

This section provides an overview of impulse radio technology andrelevant aspects of communications theory. It is provided to assist thereader with understanding the present invention and should not be usedto limit the scope of the present invention. It should be understoodthat the terminology ‘impulse radio’ is used primarily for historicalconvenience and that the terminology can be generally interchanged withthe terminology ‘impulse communications system, ultra-wideband system,or ultra-wideband communication systems’. Furthermore, it should beunderstood that the described impulse radio technology is generallyapplicable to various other impulse system applications including butnot limited to impulse radar systems and impulse positioning systems.Accordingly, the terminology ‘impulse radio’ can be generallyinterchanged with the terminology ‘impulse transmission system andimpulse reception system.’

Impulse radio refers to a radio system based on short, wide bandwidthpulses. An ideal impulse radio waveform is a short Gaussian monocycle.As the name suggests, this waveform attempts to approach one cycle ofradio frequency (RF) energy at a desired center frequency. Due toimplementation and other spectral limitations, this waveform may bealtered significantly in practice for a given application. Manywaveforms having very broad, or wide, spectral bandwidth approximate aGaussian shape to a useful degree.

Impulse radio can use many types of modulation, including amplitudemodulation, phase modulation, frequency modulation (including frequencyshape and wave shape modulation), time-shift modulation (also referredto as pulse-position modulation or pulse-interval modulation) and M-aryversions of these. In this document, the time-shift modulation method isoften used as an illustrative example. However, someone skilled in theart will recognize that alternative modulation approaches may, in someinstances, be used instead of or in combination with the time-shiftmodulation approach.

In impulse radio communications, inter-pulse spacing may be heldconstant or may be varied on a pulse-by-pulse basis by information, acode, or both. Generally, conventional spread spectrum systems employcodes to spread the normally narrow band information signal over arelatively wide band of frequencies. A conventional spread spectrumreceiver correlates these signals to retrieve the original informationsignal. In impulse radio communications, codes are not typically usedfor energy spreading because the monocycle pulses themselves have aninherently wide bandwidth. Codes are more commonly used forchannelization, energy smoothing in the frequency domain, resistance tointerference, and reducing the interference potential to nearbyreceivers. Such codes are commonly referred to as time-hopping codes orpseudo-noise (PN) codes since their use typically causes inter-pulsespacing to have a seemingly random nature. PN codes may be generated bytechniques other than pseudorandom code generation. Additionally, pulsetrains having constant, or uniform, pulse spacing are commonly referredto as uncoded pulse trains. A pulse train with uniform pulse spacing,however, may be described by a code that specifies non-temporal, i.e.,non-time related, pulse characteristics.

In impulse radio communications utilizing time-shift modulation,information comprising one or more bits of data typically time-positionmodulates a sequence of pulses. This yields a modulated, coded timingsignal that comprises a train of pulses from which a typical impulseradio receiver employing the same code may demodulate and, if necessary,coherently integrate pulses to recover the transmitted information.

The impulse radio receiver is typically a direct conversion receiverwith a cross correlator front-end that coherently converts anelectromagnetic pulse train of monocycle pulses to a baseband signal ina single stage. The baseband signal is the basic information signal forthe impulse radio communications system. A subcarrier may also beincluded with the baseband signal to reduce the effects of amplifierdrift and low frequency noise. Typically, the subcarrier alternatelyreverses modulation according to a known pattern at a rate faster thanthe data rate. This same pattern is used to reverse the process andrestore the original data pattern just before detection. This methodpermits alternating current (AC) coupling of stages, or equivalentsignal processing, to eliminate direct current (DC) drift and errorsfrom the detection process. This method is described in more detail inU.S. Pat. No. 5,677,927 to Fullerton et al.

Waveforms

Impulse transmission systems are based on short, wide band pulses.Different pulse waveforms, or pulse types, may be employed toaccommodate requirements of various applications. Typical ideal pulsetypes used in analysis include a Gaussian pulse doublet (also referredto as a Gaussian monocycle), pulse triplet, and pulse quadlet asdepicted in FIGS. 1A through 1D. An actual received waveform thatclosely resembles the theoretical pulse quadlet is shown in FIG. 1E. Apulse type may also be a wavelet set produced by combining two or morepulse waveforms (e.g., a doublet/triplet wavelet set), or families oforthogonal wavelets. Additional pulse designs include chirped pulses andpulses with multiple zero crossings, or bursts of cycles. Thesedifferent pulse types may be produced by methods described in the patentdocuments referenced above or by other methods understood by one skilledin the art.

For analysis purposes, it is convenient to model pulse waveforms in anideal manner. For example, the transmitted waveform produced bysupplying a step function into an ultra-wideband antenna may be modeledas a Gaussian monocycle. A Gaussian monocycle (normalized to a peakvalue of 1) may be described by:

${f_{mono}(t)} = {\sqrt{e}\left( \frac{t}{\sigma} \right){\mathbb{e}}^{\frac{- t^{2}}{2\sigma^{2}}}}$where σ is a time scaling parameter, t is time, and e is the naturallogarithm base.

FIG. 1F shows the power spectral density of the Gaussian pulse, doublet,triplet, and quadlet normalized to a peak density of 1. The normalizeddoublet (monocycle) is as follows:F _(mono)(f)=j(2π)√{square root over (e)}σfe ^(−2(πσf)) ²where F_(mono)( ) is the Fourier transform of f_(mono)( ), f isfrequency, and j is the imaginary unit. The center frequency (f_(c)), orfrequency of peak spectral density, of the Gaussian monocycle is:

$f_{c} = \frac{1}{2\pi\;\sigma}$

Pulse Trains

Impulse transmission systems may communicate one or more data bits witha single pulse; however, typically each data bit is communicated using asequence of pulses, known as a pulse train. As described in detail inthe following example system, the impulse radio transmitter produces andoutputs a train of pulses for each bit of information. FIGS. 2A and 2Bare illustrations of the output of a typical 10 megapulses per second(Mpps) system with uncoded, unmodulated pulses, each having a width of0.5 nanoseconds (ns). FIG. 2A shows a time domain representation of thepulse train output. FIG. 2B illustrates that the result of the pulsetrain in the frequency domain is to produce a spectrum comprising a setof comb lines spaced at the frequency of the 10 Mpps pulse repetitionrate. When the full spectrum is shown, as in FIG. 2C, the envelope ofthe comb line spectrum corresponds to the curve of the single Gaussianmonocycle spectrum in FIG. 1F. For this simple uncoded case, the powerof the pulse train is spread among roughly two hundred comb lines. Eachcomb line thus has a small fraction of the total power and presents muchless of an interference problem to a receiver sharing the band. It canalso be observed from FIG. 2A that impulse transmission systems may havevery low average duty cycles, resulting in average power lower than peakpower. The duty cycle of the signal in FIG. 2A is 0.5%, based on a 0.5ns pulse duration in a 100 ns interval.

The signal of an uncoded, unmodulated pulse train may be expressed:

${s(t)} = {a{\sum\limits_{i = 1}^{n}\;{w\left( {{c\left( {t - {iT}_{f}} \right)},b} \right)}}}$where i is the index of a pulse within a pulse train of n pulses, a ispulse amplitude, b is pulse type, c is a pulse width scaling parameter,w(t, b) is the normalized pulse waveform, and T_(f) is pulse repetitiontime, also referred to as frame time.

The Fourier transform of a pulse train signal over a frequency bandwidthof interest may be determined by summing the phasors of the pulses foreach code time shift, and multiplying by the Fourier transform of thepulse function:

${S(f)} = {a{{\sum\limits_{i = 1}^{n}\;{\mathbb{e}}^{{- j}\; 2\pi\;{fiT}_{f}}}}{W(f)}}$where S(f) is the amplitude of the spectral response at a givenfrequency, f is the frequency being analyzed, T_(f) is the relative timedelay of each pulse from the start of time period, W(f) is the Fouriertransform of the pulse, w(t,b), and n is the total number of pulses inthe pulse train.

A pulse train can also be characterized by its autocorrelation andcross-correlation properties. Autocorrelation properties pertain to thenumber of pulse coincidences (i.e., simultaneous arrival of pulses) thatoccur when a pulse train is correlated against an instance of itselfthat is offset in time. Of primary importance is the ratio of the numberof pulses in the pulse train to the maximum number of coincidences thatoccur for any time offset across the period of the pulse train. Thisratio is commonly referred to as the main-lobe-to-peak-side-lobe ratio,where the greater the ratio, the easier it is to acquire and track asignal.

Cross-correlation properties involve the potential for pulses from twodifferent signals simultaneously arriving, or coinciding, at a receiver.Of primary importance are the maximum and average numbers of pulsecoincidences that may occur between two pulse trains. As the number ofcoincidences increases, the propensity for data errors increases.Accordingly, pulse train cross-correlation properties are used indetermining channelization capabilities of impulse transmission systems(i.e., the ability to simultaneously operate within close proximity).

Coding

Specialized coding techniques can be employed to specify temporal and/ornon-temporal pulse characteristics to produce a pulse train havingcertain spectral and/or correlation properties. For example, byemploying a Pseudo-Noise (PN) code to vary inter-pulse spacing, theenergy in the uncoded comb lines presented in FIGS. 2B and 2C can bedistributed to other frequencies as depicted in FIG. 2D, therebydecreasing the peak spectral density within a bandwidth of interest.Note that the spectrum retains certain properties that depend on thespecific (temporal) PN code used. Spectral properties can be similarlyaffected by using non-temporal coding (e.g., inverting certain pulses).

Coding provides a method of establishing independent communicationchannels. Specifically, families of codes can be designed such that thenumber of pulse coincidences between pulse trains produced by any twocodes will be minimal. For example, FIG. 3 depicts cross-correlationproperties of two codes that have no more than four coincidences for anytime offset. Generally, keeping the number of pulse collisions minimalrepresents a substantial attenuation of the unwanted signal.

Coding can also be used to facilitate signal acquisition. For example,coding techniques can be used to produce pulse trains with a desirablemain-lobe-to-side-lobe ratio. In addition, coding can be used to reduceacquisition algorithm search space.

Coding methods for specifying temporal and non-temporal pulsecharacteristics are described in commonly owned, co-pending applicationstitled “A Method and Apparatus for Positioning Pulses in Time,”application Ser. No. 09/592,249, and “A Method for SpecifyingNon-Temporal Pulse Characteristics,” application Ser. No. 09/592,250,both filed Jun. 12, 2000, and both of which are incorporated herein byreference.

Typically, a code consists of a number of code elements having integeror floating-point values. A code element value may specify a singlepulse characteristic or may be subdivided into multiple components, eachspecifying a different pulse characteristic. Code element or codecomponent values typically map to a pulse characteristic value layoutthat may be fixed or non-fixed and may involve value ranges, discretevalues, or a combination of value ranges and discrete values. A valuerange layout specifies a range of values that is divided into componentsthat are each subdivided into subcomponents, which can be furthersubdivided, as desired. In contrast, a discrete value layout involvesuniformly or non-uniformly distributed discrete values. A non-fixedlayout (also referred to as a delta layout) involves delta valuesrelative to some reference value. Fixed and non-fixed layouts, andapproaches for mapping code element/component values, are described inco-owned, co-pending applications, titled “Method for Specifying PulseCharacteristics using Codes,” application Ser. No. 09/592,290 and “AMethod and Apparatus for Mapping Pulses to a Non-Fixed Layout,”application Ser. No. 09/591,691, both filed on Jun. 12, 2000, both ofwhich are incorporated herein by reference.

A fixed or non-fixed characteristic value layout may include anon-allowable region within which a pulse characteristic value isdisallowed. A method for specifying non-allowable regions is describedin co-owned, co-pending application titled “A Method for SpecifyingNon-Allowable Pulse Characteristics,” application Ser. No. 09/592,289,filed Jun. 12, 2000, and incorporated herein by reference. A relatedmethod that conditionally positions pulses depending on whether codeelements map to non-allowable regions is described in co-owned,co-pending application, titled “A Method and Apparatus for PositioningPulses Using a Layout having Non-Allowable Regions,” application Ser.No. 09/592,248 filed Jun. 12, 2000, and incorporated herein byreference.

The signal of a coded pulse train can be generally expressed by:

${s_{tr}(t)} = {\sum\limits_{i}\;{\left( {- 1} \right)^{f_{i}}a_{i}{w\left( {{c_{i}\left( {t - T_{i}} \right)},b_{i}} \right)}}}$where s_(tr)(t) is the coded pulse train signal, i is the index of apulse within the pulse train, (−1)^(fi), a_(i), b_(i), c_(i), andω(t,b_(i)) are the coded polarity, pulse amplitude, pulse type, pulsewidth, and normalized pulse waveform of the i^(th) pulse, and T_(i) isthe coded time shift of the i^(th) pulse.

Various numerical code generation methods can be employed to producecodes having certain correlation and spectral properties. Detaileddescriptions of numerical code generation techniques are included in aco-owned, co-pending patent application titled “A Method and Apparatusfor Positioning Pulses in Time,” application Ser. No. 09/592,248, filedJun. 12, 2000, and incorporated herein by reference.

It may be necessary to apply predefined criteria to determine whether agenerated code, code family, or a subset of a code is acceptable for usewith a given UWB application. Criteria may include correlationproperties, spectral properties, code length, non-allowable regions,number of code family members, or other pulse characteristics. A methodfor applying predefined criteria to codes is described in co-owned,co-pending application, titled “A Method and Apparatus for SpecifyingPulse Characteristics using a Code that Satisfies Predefined Criteria,”application Ser. No. 09/592,288, filed Jun. 12, 2000, and incorporatedherein by reference.

In some applications, it may be desirable to employ a combination ofcodes. Codes may be combined sequentially, nested, or sequentiallynested, and code combinations may be repeated. Sequential codecombinations typically involve switching from one code to the next afterthe occurrence of some event and may also be used to support multicastcommunications. Nested code combinations may be employed to producepulse trains having desirable correlation and spectral properties. Forexample, a designed code may be used to specify value range componentswithin a layout and a nested pseudorandom code may be used to randomlyposition pulses within the value range components. With this approach,correlation properties of the designed code are maintained since thepulse positions specified by the nested code reside within the valuerange components specified by the designed code, while the randompositioning of the pulses within the components results in particularspectral properties. A method for applying code combinations isdescribed in co-owned, co-pending application, titled “A Method andApparatus for Applying Codes Having Pre-Defined Properties,” applicationSer. No. 09/591,690, filed Jun. 12, 2000, and incorporated herein byreference.

Modulation

Various aspects of a pulse waveform may be modulated to conveyinformation and to further minimize structure in the resulting spectrum.Amplitude modulation, phase modulation, frequency modulation, time-shiftmodulation and M-ary versions of these were proposed in U.S. Pat. No.5,677,927 to Fullerton et al., previously incorporated by reference.Time-shift modulation can be described as shifting the position of apulse either forward or backward in time relative to a nominal coded (oruncoded) time position in response to an information signal. Thus, eachpulse in a train of pulses is typically delayed a different amount fromits respective time base clock position by an individual code delayamount plus a modulation time shift. This modulation time shift isnormally very small relative to the code shift. In a 10 Mpps system witha center frequency of 2 GHz, for example, the code may command pulseposition variations over a range of 100 ns, whereas, the informationmodulation may shift the pulse position by 150 ps. This two-state‘early-late’ form of time shift modulation is depicted in FIG. 4A.

A generalized expression for a pulse train with ‘early-late’ time-shiftmodulation over a data symbol time is:

${s_{tr}(t)} = {\sum\limits_{i = 1}^{N_{s}}\;{\left( {- 1} \right)^{f_{i}}a_{i}{w\left( {{c_{i}\left( {t - T_{i} - {\delta\; d_{k}}} \right)},b_{i}} \right)}}}$where k is the index of a data symbol (e.g., bit), i is the index of apulse within the data symbol, N_(s) is the number of pulses per symbol,(−1)^(fi) is a coded polarity (flipping) pattern (sequence), a_(i) is acoded amplitude pattern, b_(i) is a coded pulse type (shape) pattern,c_(i) is a coded pulse width pattern, and w(t,b_(i)) is a normalizedpulse waveform of the i^(th) pulse, T_(i) is the coded time shift of thei^(th) pulse, δ is the time shift added when the transmitted symbol is 1(instead of 0), d_(k) is the data (i.e., 0 or 1) transmitted by thetransmitter. In this example, the data value is held constant over thesymbol interval. Similar expressions can be derived to accommodate otherproposed forms of modulation.

An alternative form of time-shift modulation can be described asOne-of-Many Position Modulation (OMPM). The OMPM approach, shown in FIG.4B, involves shifting a pulse to one of N possible modulation positionsabout a nominal coded (or uncoded) time position in response to aninformation signal, where N represents the number of possible states.For example, if N were four (4), two data bits of information could beconveyed. For further details regarding OMPM, see “Apparatus, System andMethod for One-of-Many Position Modulation in an Impulse RadioCommunication System,” application Ser. No. 09/875,290, filed Jun. 7,2001, assigned to the assignee of the present invention, andincorporated herein by reference.

An impulse radio communications system can employ flip modulationtechniques to convey information. The simplest flip modulation techniqueinvolves transmission of a pulse or an inverted (or flipped) pulse torepresent a data bit of information, as depicted in FIG. 4C. Flipmodulation techniques may also be combined with time-shift modulationtechniques to create two, four, or more different data states. One suchflip with shift modulation technique is referred to as Quadrature FlipTime Modulation (QFTM). The QFTM approach is illustrated in FIG. 4D.Flip modulation techniques are further described in patent applicationtitled “Apparatus, System and Method for Flip Modulation in an ImpulseRadio Communication System,” application Ser. No. 09/537,692, filed Mar.29, 2000, assigned to the assignee of the present invention, andincorporated herein by reference.

Vector modulation techniques may also be used to convey information.Vector modulation includes the steps of generating and transmitting aseries of time-modulated pulses, each pulse delayed by one of at leastfour pre-determined time delay periods and representative of at leasttwo data bits of information, and receiving and demodulating the seriesof time-modulated pulses to estimate the data bits associated with eachpulse. Vector modulation is shown in FIG. 4E. Vector modulationtechniques are further described in patent application titled “VectorModulation System and Method for Wideband Impulse Radio Communications,”application Ser. No. 09/169,765, filed Dec. 9, 1999, assigned to theassignee of the present invention, and incorporated herein by reference.

Reception and Demodulation

Impulse radio systems operating within close proximity to each other maycause mutual interference. While coding minimizes mutual interference,the probability of pulse collisions increases as the number ofcoexisting impulse radio systems rises. Additionally, various othersignals may be present that cause interference. Impulse radios canoperate in the presence of mutual interference and other interferingsignals, in part because they typically do not depend on receiving everytransmitted pulse. Except for single pulse per bit systems, impulseradio receivers perform a correlating, synchronous receiving function(at the RF level) that uses sampling and combining, or integration, ofmany pulses to recover transmitted information. Typically, 1 to 1000 ormore pulses are integrated to yield a single data bit thus diminishingthe impact of individual pulse collisions, where the number of pulsesthat must be integrated to successfully recover transmitted informationdepends on a number of variables including pulse rate, bit rate, rangeand interference levels.

Interference Resistance

Besides providing channelization and energy smoothing, coding makesimpulse radios highly resistant to interference by enablingdiscrimination between intended impulse transmissions and interferingtransmissions. This property is desirable since impulse radio systemsmust share the energy spectrum with conventional radio systems and withother impulse radio systems.

FIG. 5A illustrates the result of a narrow band sinusoidal interferencesignal 502 overlaying an impulse radio signal 504. At the impulse radioreceiver, the input to the cross correlation would include the narrowband signal 502 and the received ultrawide-band impulse radio signal504. The input is sampled by the cross correlator using a templatesignal 506 positioned in accordance with a code. Without coding, thecross correlation would sample the interfering signal 502 with suchregularity that the interfering signals could cause interference to theimpulse radio receiver. However, when the transmitted impulse signal iscoded and the impulse radio receiver template signal 506 is synchronizedusing the identical code, the receiver samples the interfering signalsnon-uniformly. The samples from the interfering signal add incoherently,increasing roughly according to the square root of the number of samplesintegrated. The impulse radio signal samples, however, add coherently,increasing directly according to the number of samples integrated. Thus,integrating over many pulses overcomes the impact of interference.

Processing Gain

Impulse radio systems have exceptional processing gain due to their widespreading bandwidth. For typical spread spectrum systems, the definitionof processing gain, which quantifies the decrease in channelinterference when wide-band communications are used, is the ratio of thebandwidth of the channel to the bit rate of the information signal. Forexample, a conventional narrow band direct sequence spread spectrumsystem with a 10 kbps data rate and a 10 MHz spread bandwidth yields aprocessing gain of 1000, or 30 dB. However, far greater processing gainsare achieved by impulse radio systems, where the same 10 kbps data rateis spread across a much greater 2 GHz spread bandwidth, resulting in atheoretical processing gain of 200,000, or 53 dB.

Capacity

It can be shown theoretically, using signal-to-noise arguments, that foran impulse radio system with an information rate of a few tens of kbps,thousands of simultaneous channels could be available as a result of itsexceptional processing gain.

The average output signal-to-noise ratio of a reference impulse radioreceiver may be calculated for randomly selected time-hopping codes as afunction of the number of active users, N_(u), as:

${S_{out}\left( N_{u} \right)} = \frac{1}{\frac{1}{S_{out}(1)} + {\frac{1}{N_{s}}\frac{\sigma_{a}^{2}}{m_{p}^{2}}{\sum\limits_{k = 2}^{N_{u}}\;\left( \frac{A_{k}}{A_{1}} \right)}}}$where N_(s) is the number of pulses integrated per bit of information,A_(l) is the received amplitude of the desired transmitter, A_(k) is thereceived amplitude of interfering transmitter k's signal at thereference receiver, and σ_(rec) ² is the variance of the receiver noisecomponent at the pulse train integrator output in the absence of aninterfering transmitter. The waveform-dependent parameters m_(p) andσ_(a) ² are given by

m_(p) = ∫_(−∞)^(∞)w(t)[w(t) − w(t − δ)]𝕕tand

σ_(a)² = T_(f)⁻¹∫_(−∞)^(∞)[∫_(−∞)^(∞)w(t − s)υ(t)𝕕t]² 𝕕s,where w(t) is the transmitted waveform, v(t)=w(t)−w(t−δ) is the templatesignal waveform, δ is the modulation time shift between a digital oneand a zero value data bit, T_(f) is the pulse repetition time, or frametime, and s is an integration parameter. The output signal to noiseratio that one might observe in the absence of interference is given by:

${S_{out}(1)} = \frac{\left( {A_{1}N_{s}m_{p}} \right)^{2}}{\sigma_{rec}^{2}}$where, σ_(rec) ² is the variance of the receiver noise component at thepulse train integrator output in the absence of an interferingtransmitter. Further details of this analysis can be found in R. A.Scholtz, “Multiple Access with Time-Hopping Impulse Modulation,” Proc.MILCOM, Boston, Mass., Oct. 11-14, 1993.

Multipath and Propagation

One of the advantages of impulse radio is its resistance to multipathfading effects. Conventional narrow band systems are subject tomultipath through the Rayleigh fading process, where the signals frommany delayed reflections combine at the receiver antenna according totheir seemingly random relative phases resulting in possible summationor possible cancellation, depending on the specific propagation to agiven location. Multipath fading effects are most adverse where a directpath signal is weak relative to multipath signals, which represents asubstantial portion of the potential coverage area of a typical radiosystem. In a mobile system, received signal strength fluctuates due tothe changing mix of multipath signals that vary as the mobile unitsposition varies relative to fixed transmitters, other mobiletransmitters and signal-reflecting surfaces in the environment.

Impulse radios, however, can be substantially resistant to multipatheffects. Impulses arriving from delayed multipath reflections typicallyarrive outside of the correlation time and, thus, may be ignored. Thisprocess is described in detail with reference to FIGS. 5B and 5C. FIG.5B illustrates a typical multipath situation, such as in a building,where there are many reflectors 504B, 505B. In this figure, atransmitter 506B transmits a signal that propagates along three paths,the direct path 501B, path 1 502B, and path2 503B, to a receiver 508B,where the multiple reflected signals are combined at the antenna. Thedirect path 501B, representing the straight-line distance between thetransmitter and receiver, is the shortest. Path 1 502B represents amultipath reflection with a distance very close to that of the directpath. Path 2 503B represents a multipath reflection with a much longerdistance. Also shown are elliptical (or, in space, ellipsoidal) tracesthat represent other possible locations for reflectors that wouldproduce paths having the same distance and thus the same time delay.

FIG. 5C illustrates the received composite pulse waveform resulting fromthe three propagation paths 501B, 502B, and 503B shown in FIG. 5B. Inthis figure, the direct path signal 501B is shown as the first pulsesignal received. The path 1 and path 2 signals 502B, 503B comprise theremaining multipath signals, or multipath response, as illustrated. Thedirect path signal is the reference signal and represents the shortestpropagation time. The path 1 signal is delayed slightly and overlaps andenhances the signal strength at this delay value. The path 2 signal isdelayed sufficiently that the waveform is completely separated from thedirect path signal. Note that the reflected waves are reversed inpolarity. If the correlator template signal is positioned such that itwill sample the direct path signal, the path 2 signal will not besampled and thus will produce no response. However, it can be seen thatthe path 1 signal has an effect on the reception of the direct pathsignal since a portion of it would also be sampled by the templatesignal. Generally, multipath signals delayed less than one quarter wave(one quarter wave is about 1.5 inches, or 3.5 cm at 2 GHz centerfrequency) may attenuate the direct path signal. This region isequivalent to the first Fresnel zone in narrow band systems. Impulseradio, however, has no further nulls in the higher Fresnel zones. Thisability to avoid the highly variable attenuation from multipath givesimpulse radio significant performance advantages.

FIGS. 5D, 5E, and 5F represent the received signal from a TM-UWBtransmitter in three different multipath environments. These figures areapproximations of typical signal plots. FIG. 5D illustrates the receivedsignal in a very low multipath environment. This may occur in a buildingwhere the receiver antenna is in the middle of a room and is arelatively short, distance, for example, one meter, from thetransmitter. This may also represent signals received from a largerdistance, such as 100 meters, in an open field where there are noobjects to produce reflections. In this situation, the predominant pulseis the first received pulse and the multipath reflections are too weakto be significant. FIG. 5E illustrates an intermediate multipathenvironment. This approximates the response from one room to the next ina building. The amplitude of the direct path signal is less than in FIG.5D and several reflected signals are of significant amplitude. FIG. 5Fapproximates the response in a severe multipath environment such aspropagation through many rooms, from corner to corner in a building,within a metal cargo hold of a ship, within a metal truck trailer, orwithin an intermodal shipping container. In this scenario, the main pathsignal is weaker than in FIG. 5E. In this situation, the direct pathsignal power is small relative to the total signal power from thereflections.

An impulse radio receiver can receive the signal and demodulate theinformation using either the direct path signal or any multipath signalpeak having sufficient signal-to-noise ratio. Thus, the impulse radioreceiver can select the strongest response from among the many arrivingsignals. In order for the multipath signals to cancel and produce a nullat a given location, dozens of reflections would have to be cancelledsimultaneously and precisely while blocking the direct path, which is ahighly unlikely scenario. This time separation of multipath signalstogether with time resolution and selection by the receiver permit atype of time diversity that virtually eliminates cancellation of thesignal. In a multiple correlator rake receiver, performance is furtherimproved by collecting the signal power from multiple signal peaks foradditional signal-to-noise performance.

In a narrow band system subject to a large number of multipathreflections within a symbol (bit) time, the received signal isessentially a sum of a large number of sine waves of random amplitudeand phase. In the idealized limit, the resulting envelope amplitude hasbeen shown to follow a Rayleigh probability density as follows:

${p(r)} = {\frac{r}{\sigma^{2}}{\exp\left( \frac{- r^{2}}{2\;\sigma^{2}} \right)}}$where r is the envelope amplitude of the combined multipath signals, and2σ² is the expected value of the envelope power of the combinedmultipath signals. The Rayleigh distribution curve in FIG. 5G shows that10% of the time, the signal is more than 10 dB attenuated. This suggeststhat a 10 dB fade margin is needed to provide 90% link reliability.Values of fade margin from 10 dB to 40 dB have been suggested forvarious narrow band systems, depending on the required reliability.Although multipath fading can be partially improved by such techniquesas antenna and frequency diversity, these techniques result inadditional complexity and cost.

In a high multipath environment such as inside homes, offices,warehouses, automobiles, trailers, shipping containers, or outside in anurban canyon or in other situations where the propagation is such thatthe received signal is primarily scattered energy, impulse radio systemscan avoid the Rayleigh fading mechanism that limits performance ofnarrow band systems, as illustrated in FIGS. 5H and 5I. FIG. 5H depictsan impulse radio system in a high multipath environment 500H consistingof a transmitter 506H and a receiver 508H. A transmitted signal followsa direct path 501H and reflects off of reflectors 503H via multiplepaths 502H. FIG. 5I illustrates the combined signal received by thereceiver 508H over time with the vertical axis being signal strength involts and the horizontal axis representing time in nanoseconds. Thedirect path 501H results in the direct path signal 5021 while themultiple paths 502H result in multipath signals 5041. UWB system canthus resolve the reflections into separate time intervals which can bereceived separately. Thus, the UWB system can select the strongest orotherwise most desirable reflection from among the numerous reflections.This yields a multipath diversity mechanism with numerous paths makingit highly resistant to Rayleigh fading. Whereas, in a narrow bandsystems, the reflections arrive within the minimum time resolution ofone bit or symbol time which results in a single vector summation of thedelayed signals with no inherent diversity.

Distance Measurement and Positioning

Impulse systems can measure distances to relatively fine resolutionbecause of the absence of ambiguous cycles in the received waveform.Narrow band systems, on the other hand, are limited to the modulationenvelope and cannot easily distinguish precisely which RF cycle isassociated with each data bit because the cycle-to-cycle amplitudedifferences are so small they are masked by link or system noise. Sincean impulse radio waveform has minimal multi-cycle ambiguity, it isfeasible to determine waveform position to less than a wavelength in thepresence of noise. This time position measurement can be used to measurepropagation delay to determine link distance to a high degree ofprecision. For example, 30 ps of time transfer resolution corresponds toapproximately centimeter distance resolution. See, for example, U.S.Pat. No. 6,133,876, issued Oct. 17, 2000, titled “System and Method forPosition Determination by Impulse Radio,” and U.S. Pat. No. 6,111,536,issued Aug. 29, 2000, titled “System and Method for Distance Measurementby Inphase and Quadrature Signals in a Radio System,” both of which areincorporated herein by reference.

In addition to the methods articulated above, impulse radio technologyin a Time Division Multiple Access (TDMA) radio system can achievegeo-positioning capabilities to high accuracy and fine resolution. Thisgeo-positioning method is described in U.S. Pat. No. 6,300,903, issuedOct. 9, 2001, titled “System and Method for Person or Object PositionLocation Utilizing Impulse Radio,” which is incorporated herein byreference.

Power Control

Power control systems comprise a first transceiver that transmits animpulse radio signal to a second transceiver. A power control update iscalculated according to a performance measurement of the signal receivedat the second transceiver. The transmitter power of either transceiver,depending on the particular setup, is adjusted according to the powercontrol update. Various performance measurements are employed tocalculate a power control update, including bit error rate,signal-to-noise ratio, and received signal strength, used alone or incombination. Interference is thereby reduced, which may improveperformance where multiple impulse radios are operating in closeproximity and their transmissions interfere with one another. Reducingthe transmitter power of each radio to a level that producessatisfactory reception increases the total number of radios that canoperate in an area without mutual interference. Reducing transmitterpower can also increase transceiver efficiency.

For greater elaboration of impulse radio power control, see patentapplication titled “System and Method for Impulse Radio Power Control,”application Ser. No. 09/332,501, filed Jun. 14, 1999, assigned to theassignee of the present invention, and incorporated herein by reference.

Exemplary Transceiver Implementation

Transmitter

An exemplary embodiment of an impulse radio transmitter 602 of animpulse radio communication system having an optional subcarrier channelwill now be described with reference to FIG. 6.

The transmitter 602 comprises a time base 604 that generates a periodictiming signal 606. The time base 604 typically comprises a voltagecontrolled oscillator (VCO), or the like, having a high timing accuracyand low jitter. The control voltage to adjust the VCO center frequencyis set at calibration to the desired center frequency used to define thetransmitter's nominal pulse repetition rate. The periodic timing signal606 is supplied to a precision timing generator 608.

The precision timing generator 608 supplies synchronizing signals 610 tothe code source 612 and utilizes the code source output 614, togetherwith an optional, internally generated subcarrier signal, and aninformation signal 616, to generate a modulated, coded timing signal618.

An information source 620 supplies the information signal 616 to theprecision timing generator 608. The information signal 616 can be anytype of intelligence, including digital bits representing voice, data,imagery, or the like, analog signals, or complex signals.

A pulse generator 622 uses the modulated, coded timing signal 618 as atrigger signal to generate output pulses. The output pulses are providedto a transmit antenna 624 via a transmission line 626 coupled thereto.The output pulses are converted into propagating electromagnetic pulsesby the transmit antenna 624. The electromagnetic pulses (also called theemitted signal) propagate to an impulse radio receiver 702, such asshown in FIG. 7, through a propagation medium. In a preferredembodiment, the emitted signal is wide-band or ultrawide-band,approaching a monocycle pulse as in FIG. 1B. However, the emitted signalmay be spectrally modified by filtering of the pulses, which may causethem to have more zero crossings (more cycles) in the time domain,requiring the radio receiver to use a similar waveform as the templatesignal for efficient conversion.

Receiver

An exemplary embodiment of an impulse radio receiver (hereinafter calledthe receiver) for the impulse radio communication system is nowdescribed with reference to FIG. 7.

The receiver 702 comprises a receive antenna 704 for receiving apropagated impulse radio signal 706. A received signal 708 is input to across correlator or sampler 710, via a receiver transmission line,coupled to the receive antenna 704. The cross correlation 710 produces abaseband output 712.

The receiver 702 also includes a precision timing generator 714, whichreceives a periodic timing signal 716 from a receiver time base 718.This time base 718 may be adjustable and controllable in time,frequency, or phase, as required by the lock loop in order to lock onthe received signal 708. The precision timing generator 714 providessynchronizing signals 720 to the code source 722 and receives a codecontrol signal 724 from the code source 722. The precision timinggenerator 714 utilizes the periodic timing signal 716 and code controlsignal 724 to produce a coded timing signal 726. The template generator728 is triggered by this coded timing signal 726 and produces a train oftemplate signal pulses 730 ideally having waveforms substantiallyequivalent to each pulse of the received signal 708. The code forreceiving a given signal is the same code utilized by the originatingtransmitter to generate the propagated signal. Thus, the timing of thetemplate pulse train matches the timing of the received signal pulsetrain, allowing the received signal 708 to be synchronously sampled inthe correlator 710. The correlator 710 preferably comprises a multiplierfollowed by a short term integrator to sum the multiplier product overthe pulse interval.

The output of the correlator 710 may be coupled to an optionalsubcarrier demodulator 732, which demodulates the subcarrier informationsignal from the optional subcarrier, when used. The purpose of theoptional subcarrier process, when used, is to move the informationsignal away from DC (zero frequency) to improve immunity to lowfrequency noise and offsets. The output of the subcarrier demodulator isthen filtered or integrated in the pulse summation stage 734. A digitalsystem embodiment is shown in FIG. 7. In this digital system, a sampleand hold 736 samples the output 735 of the pulse summation stage 734synchronously with the completion of the summation of a digital bit orsymbol. The output of sample and hold 736 is then compared with anominal zero (or reference) signal output in a detector stage 738 toprovide an output signal 739 representing the digital state of theoutput voltage of sample and hold 736.

The baseband signal 712 is also input to a lowpass filter 742 (alsoreferred to as lock loop filter 742). A control loop comprising thelowpass filter 742, time base 718, precision timing generator 714,template generator 728, and correlator 710 is used to maintain propertiming between the received signal 708 and the template. The loop errorsignal 744 is processed by the loop filter to provide adjustments to theadjustable time base 718 to correct the relative time position. of theperiodic timing signal 726 for best reception of the received signal708.

In a transceiver embodiment, substantial economy can be achieved bysharing part or all of several of the functions of the transmitter 602and receiver 702. Some of these include the time base 718, precisiontiming generator 714, code source 722, antenna 704, and the like.

FIGS. 8A-8C illustrate the cross correlation process and the correlationfunction. FIG. 8A shows the waveform of a template signal. FIG. 8B showsthe waveform of a received impulse radio signal at a set of severalpossible time offsets. FIG. 8C represents the output of the crosscorrelator for each of the time offsets of FIG. 8B. For any given pulsereceived, there is a corresponding point that is applicable on thisgraph. This is the point corresponding to the time offset of thetemplate signal used to receive that pulse. Further examples and detailsof precision timing can be found described in U.S. Pat. No. 5,677,927and U.S. Pat. No. 6,304,623, issued Oct. 16, 2001, titled “PrecisionTiming Generator System and Method,” both of which are incorporatedherein by reference.

Because of the unique nature of impulse radio receivers, severalmodifications have been recently made to enhance system capabilities.Modifications include the utilization of multiple correlators to measurethe impulse response of a channel to the maximum communications range ofthe system and to capture information on data symbol statistics.Further, multiple correlators enable rake pulse correlation techniques,more efficient acquisition and tracking implementations, variousmodulation schemes, and collection of time-calibrated pictures ofreceived waveforms. For greater elaboration of multiple correlatortechniques, see patent application titled “System and Method of usingMultiple Correlator Receivers in an Impulse Radio System”, applicationSer. No. 09/537,264, filed Mar. 29, 2000, assigned to the assignee ofthe present invention, and incorporated herein by reference.

Methods to improve the speed at which a receiver can acquire and lockonto an incoming impulse radio signal have been developed. In oneapproach, a receiver includes an adjustable time base to output asliding periodic timing signal having an adjustable repetition rate anda decode timing modulator to output a decode signal in response to theperiodic timing signal. The impulse radio signal is cross-correlatedwith the decode signal to output a baseband signal. The receiverintegrates T samples of the baseband signal and a threshold detectoruses the integration results to detect channel coincidence. A receivercontroller stops sliding the time base when channel coincidence isdetected. A counter and extra count logic, coupled to the controller,are configured to increment or decrement the address counter by one ormore extra counts after each T pulses is reached in order to shift thecode modulo for proper phase alignment of the periodic timing signal andthe received impulse radio signal. This method is described in moredetail in U.S. Pat. No. 5,832,035 to Fullerton, incorporated herein byreference.

In another approach, a receiver obtains a template pulse train and areceived impulse radio signal. The receiver compares the template pulsetrain and the received impulse radio signal. The system performs athreshold check on the comparison result. If the comparison resultpasses the threshold check, the system locks on the received impulseradio signal. The system may also perform a quick check, asynchronization check, and/or a command check of the impulse radiosignal. For greater elaboration of this approach, see the patentapplication titled “Method and System for Fast Acquisition of UltraWideband Signals,” application Ser. No. 09/538,292, filed Mar. 29, 2000,assigned to the assignee of the present invention, and incorporatedherein by reference.

A receiver has been developed that includes a baseband signal converterdevice and combines multiple converter circuits and an RF amplifier in asingle integrated circuit package. For greater elaboration of thisreceiver, see U.S. Pat. No. 6,421,389, issued Jul. 16, 2002, titled“Baseband Signal Converter for a Wideband Impulse Radio Receiver,”assigned to the assignee of the present invention, and incorporatedherein by reference

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention is directed towards detecting an impulse radiosignal, and more particularly, to fast lock and acquisition of animpulse radio signal.

An impulse radio signal is assumed to be sent by a transmitter to areceiver, which may be part of a one-way or two-way communicationsystem. In a preferred embodiment, the transmitter initially sends anacquisition data signal to the receiver to assist the receiver inacquiring and locking on the signal to be sent by the transmitter. Theacquisition data signal may comprise one or more bits and is furtherdiscussed below with respect to FIG. 15A. Each bit of the acquisitiondata sent by the transmitter typically comprises one or more pulses. Thenumber of bits and the number of pulses per bit is determined by anumber of variables, including pulse rate, bit rate, interferencelevels, range, and noise. The relative locations in time, polarity, andamplitude (also known as the code) of the pulses that comprise a bit ofthe acquisition data sent by the transmitter are predetermined and arestored in the receiver. The system may use any of a number of codingtechniques including time position coding, polarity coding, amplitudecoding or combinations of these or other methods known in the art. Forpurposes of this description, time position coding will be illustrated,but it should be apparent to one skilled in the art from the teachingherein how to apply other methods of coding to the invention. Inaddition to the pulse code, the receiver stores data for verifying theacquisition data. Hence, the receiver knows the time locations of thepulses for a bit of data relative to the other pulses in the bit, butthe receiver does not know when in time the bit begins or when in timethe acquisition data begins. For example, the receiver knows thepseudorandom sequence being used by the transmitter but does not knowthe location in time of the pseudorandom sequence. The present inventionsolves this problem by detecting the beginning of a bit, detecting thebeginning of an acquisition data stream containing the bit, andverifying the contents of the acquisition data.

The Acquisition Process

FIG. 9 illustrates a flow diagram for fast lock and acquisition of animpulse radio signal using the invention. In block 1, the fast lock andacquisition of the impulse radio signal starts. In block 2, a templatepulse train is obtained and shifted as per a scanning process. Thetemplate pulse train includes a series of pulses and is compared inblock 3 by a cross-correlator to a received impulse radio signal, whichmay or may not match the pattern of pulses in the template pulse trainbecause of the unknown time shift. Through shifting the template pulsetrain, the template pulse train is placed at various locations in timeas compared to the received impulse radio signal until a match isobtained between the template pulse train and the received impulse radiosignal, where a match is determined based on one or more acceptancecriteria.

FIG. 10A illustrates two exemplary forms of a received impulse radiosignal. The first form of a received impulse radio signal 1002 exhibitslittle, if any, multipath fading effects and as such includes directpath parts 1004 that closely resemble a corresponding transmittedimpulse radio signal. In contrast, the second form of a received impulseradio signal 1006 exhibits multipath fading effects and as such includesdirect path parts 1004 and multipath reflection parts 1008 thatcorrespond to the transmitted impulse radio signal. The length andnumber of the multipath parts 1008 vary depending on the environmentbetween a transmitter and receiver.

FIG. 10B illustrates an exemplary template pulse train 20 based on timeposition coding. The template pulse train has a length of approximatelyone bit of an impulse radio signal. Alternatively, the template pulsetrain can have a length of greater than or less than approximately onebit of an impulse radio signal. The length of a bit for an acquisitionand lock period can be the same or different than a length of a bit fora communication period after the acquisition and lock period. Theexemplary template pulse train 20 is divided into n frames 21-1, 21-2,21-3, . . . , 21-n. Each frame 21-1, 21-2, 21-3, . . . , 21-n includes asingle pulse 22-1, 22-2, 22-3, . . . , 22-n, respectively.

As an example, if each pulse 22-1, 22-2, 22-3, . . . , 22-n has a widthof approximately 0.5 ns, the template pulse train has 100 frames, andeach frame has a length of approximately 100 ns, the template pulsetrain has a total length of approximately 10 μs.

As mentioned above, the fast lock and acquisition of a received impulseradio signal 1002 or 1006 starts by generating a template pulse train 20and comparing this template pulse train to the received impulse radiosignal 1002 or 1006. If the template pulse train 20 and received impulseradio signal 1002 or 1006 match each other, then the receiver can lockonto a direct path part 1004 or one of the multipath reflection parts1008 of the received impulse radio signal 1002 or 1006. If they do notmatch each other then the template pulse train 20 is continually shiftedand compared to the received impulse radio signal 1002 or 1006 untilthere is a match between one of the shifted template pulse trains andthe received impulse radio signal 1002 or 1006. The match does not haveto be exact but should be close enough to one another such that itpasses acceptance criteria, for example, meeting or exceeding somepredetermined threshold. The operation of block 2 which includes variousways of shifting the template pulse train is described below withreference to FIGS. 11A through 11P.

FIG. 10C illustrates shifting a template pulse train 20 by an offset23-1, 23-2 or 23-3. After being located at a first position 20-1, theflow of FIG. 9 proceeds from block 2 to block 3. After the flow of FIG.9 returns to block 2 from block 6, 9, 11, or 13, the template pulsetrain 20 is offset in time by a first offset 23-1 to a second position20-2. After the flow of FIG. 9 again returns to block 2 from block 6, 9,11, or 13, the template pulse train 20 is offset in time by a secondoffset 23-2 to a third position 20-3. After the flow of FIG. 9 onceagain returns to block 2 from block 6, 9, 11, or 13, the template pulsetrain 20 is offset in time by a third offset 23-3, and so on.

FIG. 10D illustrates the information as in FIG. 10C in a cyclic format.The cyclic format shown in FIG. 10D displays each successive line insynchronization with a time reference keeping time at the system pulsetrain rate. Thus, without any offset, each successive pulse train willline up immediately beneath the preceding pulse train. Using this formatthe time offset is easily visualized relative to an uninterrupted set oftemplate pulse trains.

Following are some exemplary mathematical expressions that can be usedto describe the position of each pulse 22-1, 22-2, 22-3 and 22-n in atemplate pulse train 20. The following expression equation is describedin conjunction with FIG. 11A, which illustrates an exemplary templatepulse train:

${S_{1}(t)} = {\sum\limits_{k}{a_{k}{W\left( {t - T_{k} - t_{b}} \right)}}}$where S_(l)(t) is the template waveform comprising a plurality of pulsewaveforms, W is a single ultra-wideband (pulse) waveform, k is a pulseindex, a_(k) is the amplitude of the k^(th) pulse, T_(k) is the timedelay between the beginning of the pulse train time interval and thek^(th) pulse, t is the receiver's clock time and t_(b) is the receivertime when the code begins.

An alternative expression is described in conjunction with FIG. 1B,which illustrates an exemplary template pulse train. This alternativeexpression is particularly applicable to systems where time can bedivided into frames of equal length with a single pulse occurring duringeach frame. For this type of system the template signal may be describedas follows:

${S_{2}(t)} = {\sum\limits_{k}^{n - 1}{a_{k}{W\left( {t - {kT}_{F} - t_{k} - t_{b}} \right)}}}$where S₂(t) is the template waveform comprising a plurality of pulsewaveforms W is the ultra-wideband (pulse) waveform, k is the pulseindex, a_(k) is the amplitude of the k^(th) pulse, t_(k) is the frameoffset, t is the receiver's clock time, T_(F) is the frame interval andt_(b) is the receiver time when the code begins (see FIG. 11B for agraphical representation of the above elements). It should be understoodthat the second equation relates to frame codes and as such is a specialcase of the first equation where T_(k)=kT_(F)+t_(k). It should also beunderstood that the first equation can be used to determine the positionof each pulse in a template pulse train that has fixed frame time ornon-fixed frame time. And, the second equation can be used to determinethe position of each pulse in a template pulse train that has a fixedframe time. Reference is made to the reverse binary scanning processdescribed below and Table 1 where the concepts embodied in theseexpressions will be used to describe certain properties of theinvention.

In order to align the receiver to a transmitted signal, the first stepis to search through a space involving the relevant unknown parametersof the signal until a match is found. In a typical system, a code andstream of data are predefined and known at the receiver so that the onlyremaining parameter to be found is the relative time offset between thetransmitter clock and the receiver clock. This time offset can be foundusing a searching or scanning process.

The Scanning Process

FIG. 11C illustrates the relative alignment of a received impulse radiosignal 1006 compared to a template pulse train that was shifted inaccordance with an exemplary scanning process. Basically, the sametemplate pulse train 20 is shifted to different locations in time andcompared to the received impulse radio signal 1006 until a match isobtained between one of the shifted template pulse trains and thereceived impulse radio signal. The different locations in time where thetemplate pulse train 20 is placed are determined by the offsetsgenerated by a scanning process. For instance, the first offset is 0,the second offset can be a ¼ wavelength from the first offset, and thethird offset can be a ¼ wavelength from the second offset and a ½wavelength from the first offset and so on.

As shown in FIG. 11C, four template pulse trains 20-1, 20-2, 20-3 and20-4 are sequentially compared to the received impulse radio signal 1006(see block 3 of FIG. 9). Each template pulse train 20-1, 20-2, 20-3 and20-4 has a pulse code pattern that is similar to the pulse code pattern(direct path parts 1104 and, if any, multipath reflection parts 1108)within the received impulse radio signal 1006. First, the comparisonbetween template pulse train 20-1 shifted by offset₁ and the receivedimpulse radio signal 1006 fails to indicate a match. Next, thecomparison between template pulse train 20-2 shifted by offset₂ and thereceived impulse radio signal 1006 fails to indicate a match. Likewise,the comparison between template pulse train 20-3 shifted by offset₃ andthe received impulse radio signal 1006 fails to indicate a match. Thereare no matches between template pulse trains 20-1, 20-2 and 20-3 and thereceived impulse radio signal 1006, because the pulses in the templatepulse trains 20-1, 20-2 and 20-3 are not aligned in time with the pulsesin the received impulse radio signal 1006 (see blocks 3-6 of FIG. 9).However, the comparison between template pulse train 20-4 shifted byoffset₄ and the received impulse radio signal 1006 indicates a match,because the pulses in the template pulse trains 20-4 are aligned in timewith the pulses in the received impulse radio signal 1006. Once there isa match, the receiver locks onto any part of the impulse radio signal1006 that was aligned with the template pulse train 20-4.

It should be understood that if the offsets are relatively small andincreasing in delay (e.g., fine step scanning process), then thereceiver is likely to lock onto the direct path 1104 or leading edge ofthe received impulse radio signal 1006. However, if the offsets arerelatively large (e.g., coarse step scanning process) then the receivermay to lock onto one of the multipath reflection parts 1108 of thereceived impulse radio signal 1006. In the event the receiver locks ontoone of the multipath reflection parts 1108 of the received impulse radiosignal 1006, the receiver can then locate the direct path 1104 bycontinuing to scan for the leading edge of the received impulse radiosignal 1006.

Various alternative scanning approaches are described below includingfine step scanning, coarse step scanning, frame step scanning, randomoffset scanning, pseudo noise offset scanning, reverse binary scanning,reverse ternary scanning, reverse n-ary scanning, mixed binary scanning,and variations on reverse binary and mixed binary patterns.

Fine Step Scanning

The fine step scanning process, in brief, involves shifting the timeoffset of a receiver time base in small increments and observing areceived signal for a period of time sufficient to identify a matchbetween the received signal and a locally generated template signal inthe presence of noise and/or interference.

The fine step search process will now be described with reference toFIG. 11D. FIG. 11D is a simplified cyclic diagram showing the relativesampling positions in a fine step scanning process. Referring to FIG.11D, the top axis 1140 represents a time axis spanning a search intervalwith eight resolution positions labeled “0”-“7” and returning to “0”.The final “0” is the same as the “0” of the next search interval. Thesearch interval may, for example, correspond to the length of a templatepulse train used to receive the desired signal. The vertical axis 1142represents successive search intervals. The triangles 1144 represent areference point, such as the first pulse, in a sampling template pulsetrain. The simplified example in FIG. 11D shows eight resolution points,but in a typical system, the number of resolution points may range froma few to many tens of thousands or more, with as many correspondingsamples.

The search begins with a beginning sample 1144, which may integrate asmany pulses as necessary to receive the signal and determine a match.These pulses will preferably comprise a code length. If a possible matchis determined, the search terminates and then the system furtherexamines the signal to determine its validity and complete theacquisition process. If no match is determined, an offset value of 1resolution bin is added to the template timing and a second sample 1146is taken. The process continues until a signal is found or the finalsample 1148 is taken at which time the process may begin again.

As shown in FIG. 11D, the step is positive, increasing in time; howeveralternatively, the step may be negative, decreasing in time.

The time step size used in fine step scanning is selected to be smallenough to guarantee that received signals of all possible time offsetswill be sampled at some point in the scan. This is balanced by a generalneed to complete the search in a reasonably short period of time. Thus,the step size is made as large as practical consistent with samplingsignals at all offsets. A Nyquist sampling argument might suggest that astep size of ½ cycle time for a sine wave at the UWB center frequencymight be an upper bound on the step size. However, one can envision asystem based on ½ cycle spacing to hit the zero crossings of a pulse andmiss the signal altogether. Accordingly, a spacing of ¼ cycle is areasonable practical nominal value for a step size. The step size valuecan be refined and optimized for a particular system using detailedanalysis or experimentation as is practiced by those skilled in the art.

The fine step scanning process takes successive fine steps through thelength of the search interval, dwelling on each step long enough todetect a desired signal if present. To illustrate the timing related tothis method, consider an example system with a center frequency of 2 Ghzand 100 ns frames. If this system is designed to integrate 100 pulses toachieve detection, and if the pulse train repeats each 100 frames, thenthe system would dwell 100×100 ns=10 μs at each step. If the systemtakes ¼ wave steps, each step would be 125 ps and it would take 10μs/125 ps=80,000 steps to complete a full scan of the pulse train, whichis in this case the search interval. At 10 μs per step this is 0.8seconds. Given a uniform probability of initial signal offset over thetime of the pulse train, the average time to detection would be half ofthe full scan time, or 0.4 seconds.

Coarse Step Scanning

Coarse step scanning is a process wherein the step size is greater thanthe resolution required for thorough searching. Whereas a fine stepsearch may step ¼ cycle or so, a coarse step scan typically stepsseveral cycles or more. A search scan using coarse steps can span thelength of a search interval in far less time than by using fine steps,but the coverage is sparse and potentially leaves unsampled gaps. Theunsampled gaps can be filled by restarting each successive scan with anadditional offset such that successive scans sample previously unsampledoffset values. This process can be continued until the search intervalis sampled to the necessary resolution.

Coarse step scanning will now be described with reference to FIG. 11E.FIG. 11E is a simplified cyclic diagram showing the relative samplingpositions in a coarse step scanning process. The axes are the same asdescribed for FIG. 11D. The coarse step scanning process begins with aninitial sample 1144 and checks for match as in the fine step process,but if no match is found, a coarse step 1150 is added, which isillustrated as two resolution bins, but may be any number including anon-integer. The process continues with successive coarse steps untilthe span is completed 1152. At this point in the process, a new scan isstarted 1154 with an additional offset 1156, which allows a new position“1” to be sampled. Further samples “3”, “5”, “7” are then takenincluding this new offset 1156 to allow unsampled resolution positions“3”,“5”,“7” to be sampled. Further scans with different offsets may betaken as necessary to achieve the final resolution necessary or until asignal match is determined.

As shown in FIG. 11E, the coarse step is positive, increasing in time;however alternatively, the coarse step may be negative, decreasing intime.

It would seem that both fine step scanning and coarse step scanningwould take the same time to scan a search interval to a givenresolution. For example, using fine step scanning, where the resolutionlimit is the step size, a scan resolution of ins over a search intervalof 1000 ns requires 1000 ns for to complete the search. However, usingcoarse step scanning, where the search resolution is finer than the stepsize, a scan resolution of 1 ns over a search interval of 1000 ns usinga step size of 10 ns requires 10 scans of 100 samples, where each scanmight begin with 1 ns additional offset, which also requires 1000 ns tocomplete the complete search. Thus, it would seem that there is nobenefit in acquisition time for coarse step scanning.

A benefit arises, however, where the signal may have substantial lengthas in a narrow band signal, medium wide band signal or an ultra widebandsignal in a multipath environment.

Further insight into this benefit may be obtained with reference to FIG.11P. FIG. 11P illustrates step size selection in a multipathenvironment. FIG. 11P depicts a signal 1006 with a length 1190 based ona threshold 1191 and contained within a search interval 1140. Thisbenefit arises because the coarse step size 1150 need only be equal toor less than the signal length 1190 in order to be assured of samplingthe desired signal 1006, if the desired signal 1006 is present. Thesignal length 1190 is the span of a signal where its strength issufficient to meet a performance threshold 1191 for detection. In thecase of an ultra wideband signal in multipath, this length may encompassa certain amount of the multipath reflection signals 1008 even thoughthese reflections may not cover the entire signal length span 1190.

Accordingly one embodiment of the invention characterizes theenvironment and establishes a coarse step size 1150 in accordance withthe measured environment. The characterization may be accomplished by areceiver or may be accomplished during the design of a receiver and thenestablished in the receiver as a predetermined value. For example, ascanning receiver could be used to characterize the environment such asis described in application Ser. No. 09/537,264 previously incorporatedby reference. Alternatively, the coarse step size 1150 value may bebased on an experimental performance optimization process. The coarsestep size value 1150 may also be a factor based on average channel delayspread or RMS delay spread, for example the factor may be one.

Further details on coarse step scanning in multipath are supplied hereinalong with a discussion of reverse binary scanning.

Frame Step Scanning

As an option, the coarse step may be equal to a frame size. This canoften result in hardware simplification since hardware already existsfor generation of frame intervals for pulse coding. If the step werealways exactly equal to a frame, the system would not scan betweenpulses. This can be overcome by skewing the receiver clock or by addingan additional offset at the beginning or end of a code length.

Random Offset Scanning

As an additional example of a shifting strategy, the offset can bedetermined through a random offset generating process. For example, arandom or pseudorandom number generator can be used to select an offsetto shift the template pulse train, where the selected offset is not anoffset previously selected by the random number generator. As an option,a random offset can be generated using oscillator drift. With thisoption, the inherent drift of the oscillator can be used to adjust theoffset for shifting the template pulse train.

Pseudo Noise Offset Scanning

As a special case of random offset scanning, a pseudorandom numbergenerator, in particular a linear feedback shift register generating amaximal length sequence, or equivalent, will generate all of the offsetswithin its resolution before it repeats (except zero, which can beincluded). For example a five bit length LFSR will generate all of theintegers from 1 to 2⁵−1=31 and will then repeat the sequence. Theabsence of a zero state will probably not be noticed in the operation ofa typical system, but the zero state can be added by forcing an extracycle with dedicated logic.

Reverse Binary Scanning

In another example of a shifting strategy, the offset can be determinedvia reverse binary scanning, which can be considered a type of coarsestep scanning process. In brief, a reverse binary scanning process usesa scan offset value that is proportional to a reverse binary counter.Thus, each scan includes an offset determined by:Offset_(i) I _(s) *M _(i)where M is a multiplier multiplying the scanned interval I_(s). Themultiplier M_(i) is given by the following sequence: 0, ½, ¼, ¾, ⅛, ⅝,⅜, ⅞, 1/16, 9/16, 5/16, 13/16, 3/16, 11/16, 7/16, 15/16, 1/32, 17/32,9/32, 25/32, 5/32, 21/32, 13/32, 29/32, 3/32, 19/32, 11/32, 27/32, 7/32,23/32, 15/32, 31/32, 1/64, etc, where M₀=0 is the initial case. Thescanning may continue until I_(s) *M _(i) is less than ¼ wavelength of apulse after the initial case. As an option, the initial case can beM₀=1.

This type of scanning is referred to as reverse binary scanning becausea binary sequence of numbers is reversed from least significant bit tomost significant bit to provide the fraction of the multiplier. Forexample, for a denominator of 8 for M, three bits are needed, and themultiplier M determined as follows in Table 1:

TABLE 1 b₁b₂b₃ b₃b₂b₁ M 000 Reverse→ 000 Obtain 0/8 = 0 001 Bits 100numerator→ for 4/8 = ½ 010 010 M from binary 2/8 = ¼ 011 110 number 6/8= ¾ 100 001 ⅛ 101 101 ⅝ 110 011 ⅜ 111 111 ⅞

The same procedure may be used to extend the resolution of the sequenceby adding more bits to obtain more fractions (i.e., multipliers M) forthe offset. Once adequate resolution is achieved to assure finding therequired signal, further resolution has little effect on acquisitionspeed performance. Thus the level of precision, if sufficient, is not acritical design parameter.

Reverse Binary scanning will now be described with reference to FIG.11F. FIG. 11F is a simplified cyclic diagram showing the relativesampling positions in a reverse binary scanning process. The axes arethe same as described for FIG. 11D. The reverse binary scanning processbegins with an initial sample 1144 and checks for match as in the finestep process, but if no match is found, an offset equal to half of thesearch interval 1160 is added. If no match is found, the next offset1162 is one quarter of the search interval 1160. The third offset isthree quarters of the search interval. This process continues samplingthe remaining positions “1”,“5”,“3”,“7” and then returns to “0” to beginthe process again. Upon completion of the reverse binary search patternexample in FIG. 11F, the search interval is sampled to the sameresolution as for the fine step and coarse step examples in FIGS. 11Dand 11E respectively.

FIG. 11G illustrates the relative timing of a received impulse radiosignal 1006 compared to a template pulse train that was shifted inaccordance with a reverse binary scanning process. Basically, the sametemplate pulse train 20 is shifted to different locations in time andcompared to the received impulse radio signal 1006 until a match isobtained between one of the shifted template pulse trains and thereceived impulse radio signal. The different locations in time where thetemplate pulse train 20 is placed are determined by the offsetsgenerated by the reverse binary scanning process (see Table 1). Forinstance, the first offset is 0, the second offset is ½ of the length ofthe template pulse train 20 from the first offset, and the third offsetis ¼ of the length of the template pulse train 20 from the first offsetand so on. In this case, the scanned interval is the full length of thetemplate pulse train.

As shown in FIG. 11G, five template pulse trains 20-1, 20-2, 20-3, 20-4and 20-5 are sequentially compared to the received impulse radio signal1006 (see block 3 of FIG. 9). Each template pulse train 20-1, 20-2,20-3, 20-4 and 20-5 has a pattern of pulses that are similar in relativetiming to the pattern of pulses (direct path parts 1104 and, if any,multipath reflection parts 1108) within the received impulse radiosignal 1006. First, the comparison between template pulse train 20-1shifted by offset₁ and the received impulse radio signal 1006 fails toindicate a match. Next, the comparison between template pulse train 20-2shifted by offset₂ and the received impulse radio signal 1006 fails toindicate a match. Likewise, the comparison between template pulse train20-3 shifted by offset₃ and received impulse radio signal 1006 fails toindicate a match. Also, the comparison between template pulse train 20-4shifted by offset₄ and the received impulse radio signal 1006 fails toindicate a match. There are no matches between template pulse trains20-1, 20-2, 20-3 and 20-4 and the received impulse radio signal 1006,because the pulses in the template pulse trains 20-1, 20-2, 20-3 and20-4 are not aligned in time with the pulses in the received impulseradio signal 1006 (see blocks 3-6 of FIG. 9). However, the comparisonbetween template pulse train 20-5 shifted by offset₅ and the receivedimpulse radio signal 1006 indicates a match, because the pulses in thetemplate pulse trains 20-5 are aligned in time with the pulses in thereceived impulse radio signal 1006. Once there is a match, the receiverlocks onto any part (most likely one of the multipath reflection parts1108) of the received impulse radio signal 1006 that was aligned withthe template pulse train 20-5.

FIG. 11G illustrates in greater detail the reverse binary scanningprocess of FIG. 11F. In FIG. 11G, the received impulse signal 1006 isshown at an arbitrary time offset relative to the receiver template.Each receiver template pulse train 20-1 through 20-5 is timed accordingto a time offset (offset₁ through offset₅) relative to a receiver cyclicreference 1171 that repeats at a predetermined interval designed to besynchronous with the received signal 1006. As shown in FIG. 11G, twocode cycles are allocated for each template code pulse train 20-1through 20-5 to avoid overlap of template pulse trains.

Since, the receiver does not know when a particular transmitter is goingto transmit an impulse radio signal, the receiver continually repeatsthe scanning process in an attempt to lock onto any part of the receivedimpulse radio signal 1006. As illustrated, the first template pulsetrain scan 20-1 does not line up with the received pulse train 1006. Thefollowing pulse train 20-2 is delayed from the reference by offset₂ andit also does not line up with the received signal 1006. This processcontinues with no match until offset₅ is used, at which point, all fourpulses line up enabling the signal to be detected.

Of course, if the template pulse train 20-1 through 20-5 has a patternof pulses that are not similar to the pattern of pulses within areceived impulse radio signal then there will not be a match. Inaddition, if interference degrades the received impulse radio signal1006 then there may not be a match during one or more scanningprocesses.

Reverse Binary Scanning Example

Referring again to FIG. 11C and the associated equation:

${S(t)} = {\sum\limits_{k = 0}^{n - 1}{a_{k}{{W\left( {t - {kT}_{F} - t_{k} - t_{b}} \right)}.}}}$

A reverse binary scanning process uses a sequence of these pulse trainswhich may be represented by the following:

${B(t)} = {\sum\limits_{j = 0}^{m - 1}{S_{j}\left( {t - {jnT}_{f} - {T_{search}M_{j}}} \right)}}$${B(t)} = {\sum\limits_{j = 0}^{m - 1}{\sum\limits_{k = 0}^{n - 1}{a_{k}{W\left( {t - {kT}_{F} - t_{k} - {jnT}_{f} - {T_{search}M_{j}}} \right)}}}}$where B(t) is a pulse train according to a reverse binary searchsequence, T_(search) is the interval to be searched, M_(j) is the offsetfraction associated with each search scan interval (S_(j)(t)subsequence), nT_(f) is the length of a pulse code subsequence S_(j)(t).Note that the pulse train beginning time t_(b) is now defined by thescan sequence, jnT_(f), and reverse binary offset, T_(search)M_(j).The following is an example of how the offsets of the reverse binaryscanning process of Table 2 can be used to determine the position ofeach impulse in a template pulse train. For this example, assume thetemplate pulse train has the following characteristics:Frame time: T_(F)=100 nsCode Length: n=120=number of pulses per bitCode interval, T_(c); Search interval, T_(search):T _(c) =T _(search)=120*100 ns=12 μsScanning resolution, resolution of M_(j): 14 bitsThus, the offset value M_(j) progresses as follows:

j M(j)(binary) M(j)(decimal) 00 . . . 0000 0.0000 . . . 00 0 00 . . .0001 0.1000 . . . 00 0.5 00 . . . 0010 0.0100 . . . 00 0.25 00 . . .0011 0.1100 . . . 00 0.75 00 . . . 0100 0.0010 . . . 00 0.125 00 . . .0101 0.1010 . . . 00 0.625 . . . . . . . . .Also assume a time position code sequence, t_(k)(t₀=23 ns, t₁=41 ns,t₂=18 ns, t₃=72 ns, t₄=95 ns . . . t₁₁₉=49 ns). In addition,t_(b)=T_(search) M(j) and for this example T_(search)=T_(c) whereT_(search) is the total search space (time interval to be searched)(typically equal to the code space, or code length in time units).Therefore, a pulse occurs at t=kT_(F)+t_(k)+t_(b). The sequence for J=0and k=0 through 119 is:T=kT _(f) +t _(k) +T _(search) M _(j)t=0+23 ns+0=23 ns (location of first template pulse)t=100 ns+41 ns+0=141 ns (location of second template pulse)t=200 ns+18 ns+0=218 ns (location of third template pulse)t=300 ns+72 ns+0=372 ns (location of fourth template pulse)t=400 ns+95 ns+0=495 ns (location of fifth template pulse)...k=119 then t=11900 ns+49 ns+0=11949 ns (location of the last templatepulse)And, at J−1 and k=0 through 119 and t_(b)=12 μs×0.5=6000 nst=0+23 ns+6000 ns=6023 ns (location of first template pulse)t=100 ns+41 ns+6000 ns=6141 ns (location of second template pulse)t=200 ns+18 ns+6000 ns=6218 ns (location of third template pulse)t=300 ns+72 ns+6000 ns=6372 ns (location of fourth template pulse)t=400 ns+95 ns+6000 ns=6495 ns (location of fifth template pulse)...k=119 thent=11900 ns+49 ns+6000 ns=17949 ns (location of the last template pulse).

Reverse Ternary and Reverse n-ary Scanning

Moreover, the reverse binary scanning can be extended to reverse n-aryscanning, where n is any desired base. For instance, one type ofscanning is known as reverse ternary scanning because a sequence of base3 numbers is reversed from least significant bit to most significant bitto provide fraction of a multiplier M. Like the reverse binary scanningprocess, the multiplier M can be used to determine the offset. Forexample, for a denominator of 27 for M, two digits are needed, and themultiplier M is determined as follows in Table 2:

TABLE 2 b₁b₂ b₂b₁ M 00 →Reverse→ 00 Obtain 0/9 01 digits 10 →numerator→3/9 02 20 for M from 6/9 10 01 ternary number 1/9 11 11 4/9 12 21 7/9 2002 2/9 21 12 5/9 22 22 8/9

The same procedure continues by adding more digits to obtain morefractions (i.e., multipliers M) for the offset.

Mixed Binary Scanning

In yet another example of a coarse shifting strategy, the offset can bedetermined via a mixed binary scanning process. This type of scanning isknown as mixed binary scanning because a binary sequence of numbers isordered according to a reverse binary scanning process and anotherbinary sequence of numbers is ordered according to a forward binaryscanning process. Like the reverse binary scanning process, a multiplierM can be used to determine the offset (e.g., offset=IO*M). For example,for a denominator of 32 for M, five bits are needed, and the multiplierM=½b₁+¼b₂+⅛b₅+ 1/16b₄+ 1/32b₃ as followed in Table 3:

TABLE 3 b₁b₂b₃b₄b₅ b₁b₂b₅b₄b₃ M j 00000 00000 0 0 01000 01000 Obtain ¼ 110000 →Reverse→bits 10000 M→ ½ 2 11000 (b₃b₄b₅₎ 11000 ¾ 3 00001 00100 ⅛4 01001 01100 ⅜ 5 10001 10100 ⅝ 6 11001 11100 ⅞ 7 00010 00010 1/16 801010 01010 5/16 9 10010 →Reverse→ 10010 Obtain 9/16 . 11010 bits(b₃b₄b₅₎ 11010 M→ 13/16 . . . . . . . . . . .

The same procedure can be changed by simply adding more bits to theforward binary scanning process (shown in Table 3 as b₁b₂) and/or byadding more bits to the reverse binary scanning process (shown in Table3 as b₃b₄b₅) to obtain more fractions (i.e., multipliers M) for theoffset. Also, the order of the forward component and the reversecomponent can be reversed e.g., b₃b₂b₁b₄b₅ or b₅b₄b₃b₁b₂. Also, notethat each sequence may be truncated i.e., it need not extend to its fullcount.

Modified Reverse Binary And Mixed Binary Patterns

Reverse binary and mixed binary search patterns may be modified byinverting selected bits, i.e., exclusive or-ing the output by a maske.g., if the second bit of a three bit reverse binary counter isinverted the count changes from 0, 4, 2, 6, 1, 5, 3, 7 to 2, 6, 0, 4, 3,7, 1, 5, which also results in a widely scattered pattern.

In another variation, the same difference sequence may be preserved byselecting alternates. Note that in the sequence 0, 4, 2, 6, 1, 5, 3, 7,the differences between successive numbers forms a first differencesequence: 4, 2, 4, 5, 4, 2, 4, 1, the last digit representing the returnfrom 7 in the first sequence to zero in the following sequence (which is8 in the first sequence). An alternate sequence formed as follows: 0, 4,6, 2, 7, 3, 5, 1 results in a second difference sequence: 4, 2, 4, 5, 4,2, 4, 1. Note that the second difference sequence equals the firstdifference sequence. A further alternate sequence: 0, 4, 6, 2, 1, 5, 7,3 has a third difference sequence: 4, 2, 4, 1, 4, 2, 4, 5.

Note also that further sequences may be derived from this process bycertain operations. For example, a ones complement, or twos complement,or reversing in order of all or part of the sequence, or adding aconstant all yield equivalent sequences. For example, beginning with thesequence 0, 4, 2, 6, 1, 5, 3, 7, the ones complement is 7, 3, 5, 1, 6,2, 4, 0, which coincidentally, is also a reversing in order of the samesequence. The twos complement is 0, 4, 6, 2, 7, 3, 5, 1, which, in thiscase results in a reversal in order of the 6 and 2 and a reversal inorder of the last four digits. Adding 1 yields 1, 5, 3, 7, 2, 6, 4,8(0). Note that the appropriate math is modulo 8 for this example sothat the “8” result is equal to “0”

A feature of these sequences is that large steps resulting from the mostsignificant bits predominate the differences and smaller steps occurless frequently. In the reverse binary case, large steps resulting inhalf of the scan interval occur half of the time and smaller stepsassociated with each smaller significant bit occur less and lessfrequently. This has a particular advantage in the presence of multipathas will be described herein.

Mixed scanning can also be extended to systems using multiple scanningprocesses where reverse binary scanning is one component of the scanningprocess. In one example, an analog slow scan can be combined with areverse binary large-scale scan. In another example, a reverse binaryscan may scan a single frame and frame-to-frame scanning can be donesequentially.

In one such alternative, the reverse binary process establishes anoffset, and each frame offset is successively added to it to produce asequence of offsets for search. For example, consider a system with aframe size of 100 ns and a code length and bit size of 4 frames. Thereverse binary process may be used to span the frame size of 100 ns andthe frame counter may be used to sequence through the frames. Thus, thecombined sequence would be: 0, 100, 200, 300, 50, 150, 250, 350, 25,125, 225, 325, 75, 175, 275, 375, 12.5, 112.5, 212.5, 312.5, and so onto the desired limit of resolution for the reverse binary part of theprocess.

Moreover, in general, the mixed binary scanning can be extended to mixedn-ary scanning, where n is any desired base.

Optimization of a Reverse N-ary Scanning Order

As previously described relative to FIG. 9, the correlation result foreach pulse in the template pulse train may be integrated to obtain asingle summation result that is compared to a threshold. This approachonly requires a single cross correlator 710 and a single pulse summationstage 734 as shown in FIG. 7. Under this arrangement, the process ofshifting from one scan position to another scan position corresponds toa delay in time between scans during which cross-correlation does notoccur. In order to speed acquisition, the order of a reverse N-aryscanning sequence may be optimized to minimize the total amount of delaytime required during the scanning process. The optimization process isbased on the recognition that scan positions at a given level of scanresolution can be reordered while retaining the benefits of the reverseN-ary scanning approach and the recognition that shifting to a backwardscan position can require a longer delay than shifting to a forward scanposition and thus should be avoided where possible.

The difference between moving to a forward scan position versus movingto a backward scan position can be seen in FIG. 11H. Referring to FIG.11H, the vertical axis represents successive code periods, where a codeperiod is defined as the amount of time required to receive a pulsetrain. The top axis represents a cyclic time axis spanning a searchinterval with eight scan positions labeled “0”-“7” and returning to “0”,where due to the repeating nature of the pulse trains, the final “0” isthe same as the “0” of the next code period.

In FIG. 11H the total time (in code periods) required to perform acomplete reverse binary scan using a scan sequence of 0, 4, 2, 6, 1, 5,3, 7 is depicted. In the figure, the first scan begins at scan position0 of the first code period (code period 0) and ends at scan position 0of the second code period (code period 1), which is indicated by thesolid line with an arrow spanning the complete time axis. The shiftingforward from scan position 0 to scan position 4 requires a delay of halfa code period, which is indicated by a dashed line. Thus, before thesecond scan begins, a code period and a half of time have elapsed. Thesecond scan begins at scan position 4 of the second code period and endsat scan position 4 of the third code period (code period 2). Theshifting backward from scan position 4 to scan position 2 requires adelay of three quarters of a code period, again indicated by a dashedline, which corresponds to the last half of the third code period andthe first quarter of the fourth code period (code period 3). Thus,before the third scan begins, four and a quarter code periods of timehave elapsed. The scanning process continues requiring a total of 11.875code periods to complete.

By reordering the scan sequence, the requirement for backward shiftingbetween scans can be reduced thereby reducing the time required tocomplete the scanning process. In one embodiment of the invention, thesequence of reverse N-ary scan positions is determined by setting eachscan position of a given scan resolution to an available scan positionthat is forward of the previous scan position. In a preferredembodiment, the sequence of reverse N-ary scan positions is determinedby setting each scan position of a given scan resolution to the closestavailable scan position that is forward of the previous scan position.With either embodiment, if a forward scan position is not available, theavailable backward scan position having the lowest index is selected.

For these optimization approaches, the reverse N-ary sequence is orderedper group of scan positions making up a given scanning resolution level,where each such group is defined as those scan positions that define alevel of scan resolution when combined with all scan positions defininghigher resolution levels. For example, given the reverse binary sequence0, 4, 2, 6, 1, 5, 3, 7, scan positions 0 and 4 correspond to the firstgroup of scan positions, which define the first scanning resolutionlevel having a resolution of a half code period. Scan positions 2 and 6correspond to the second group of scan positions that, when combined toscan positions 0 and 4, define the second scanning resolution levelhaving a resolution of a quarter code period. Scan positions 1, 5, 3,and 7 correspond to the third group of scan positions that, whencombined with scan positions 0, 4, 2 and 6, define the third scanningresolution level having a resolution of an eighth of a code period. Iffiner scanning resolution is required to achieve acquisition, the nextgroup of scan positions would be made up of the scan positions 0.5, 4.5,2.5, 6.5, 1.5, 5.5, 3.5, and 7.5 that, when combined with the previousscan positions, define the fourth resolution level having a resolutionof one sixteenth of a code period, and so forth and so on for whateverlevels of scanning resolution are required to achieve acquisition.

The optimized reverse binary scan position ordering process can bevisualized in relation to FIG. 11I, which is exactly like FIG. 11Hexcept with different ordering of the scan positions. Continuing withthe example, ordering is first determined for the first scan resolutionlevel consisting of scan positions 0 and 4. The scan order wouldtypically begin with scan position 0 in which case the next scanposition would be scan position 4, which is the only available scanposition at the first scan resolution level. Although, scan position 4could be selected as first scan position if an initial one half codeperiod delay were desired, for this example, it is assumed that such adelay is not desired. Thus, the order of the scanning positions makingup the first scanning level is determined to be 0,4.

The ordering for scan positions 2 and 6 corresponding to the secondresolution level would then be determined. The only available forwardscan position from scan position 4 is scan position 6. Thus, scanposition 6 is determined to be the next scan position. Afterwards, noforward scan positions of the second resolution level are available fromscan position 6, in which case the next scan position is determined tobe scan position 2. Thus, the order of the sequence including the scanpositions of the first and second scanning resolution levels would be 0,4, 6, 2.

Next, the ordering for scan positions 1, 5, 3, and 7 corresponding tothe third scanning resolution level would be determined. From scanposition 2, three forward positions are possible 3, 5, and 7. Under oneembodiment, any of the three forward positions could be selected andhave a lesser delay than scan position 1, which would require a backwardshift. Under a preferred embodiment, scan position 3, which is theclosest of the available forward scan positions, is selected.Afterwards, the closest of the available forward scan positions is scanposition 5, so it is selected as the next scan position. Then scanposition 7 is the only remaining forward scan position, so it isselected, leaving no available forward scan positions. Because noforward scan positions remain available, the available backward scanposition having the smallest index is selected, which is scanposition 1. This optimized reverse binary scan position ordering processwould continue for each additional scanning resolution level asrequired. Thus, according to the preferred embodiment, the optimizedreverse binary sequence is 0, 4, 6, 2, 3, 5, 7, 1. The optimizedsequence requires 10.125 code periods to complete, which is a nearly 15%acquisition speed improvement over the non-optimized reverse binary scansequence 0, 4, 2, 6, 1, 5, 3, 7.

Reverse N-ary Scanning By Multiple Receiving Blocks

In one embodiment of the invention, two or more receiving blocks, eachreceiving block comprising all the elements of FIG. 7 other than theantenna 704, are coupled to the received signal 708. The two or morereceiving blocks are coupled to one or more processing blocks thatcontrol the scan process and perform processing of the two or morereceived signals 739 relative to acceptance criteria including thethreshold check, quick check, etc. in accordance with the invention.Under this arrangement, the two or more receiving blocks collaborativelyperform a reverse N-ary scan process by operating on alternate scans sothat integration intervals may overlap without interference. Alternativearrangements for employing multiple cross-correlators, which areapplicable to this embodiment of the invention, are described inapplication Ser. No. 09/537,264 previously incorporated by reference.

A timing diagram of a collaborative reverse binary scan as might beperformed by two receiving blocks is provided in FIG. 11J. Referring toFIG. 11J, the vertical axis represents successive code periods, where acode period is defined as the amount of time required to receive a pulsetrain. The top axis represents a cyclic time axis spanning a searchinterval with eight scan positions labeled “0”-“7” and returning to “0”,where due to the repeating nature of the pulse trains, the final “0” isthe same as the “0” of the next code period. For each code period oftime, the scanning activity performed by a first receiving block isdepicted by a thick solid or dashed line and the scanning activityperformed by a second receiving block is depicted by a thin solid ordashed line, where a solid line indicates a scan and a dashed lineindicates a delay. The thick and thin lines (solid or dashed) are alsobracketed to provide further indication that the depicted scanningactivity by the two receiving blocks occurs simultaneously during thegiven code period.

In FIG. 11J the total time (in code periods) required by two receivingblocks to perform a complete reverse binary scan using a scan sequenceof 0, 4, 2, 6, 1, 5, 3, 7 is depicted. In the figure, the first scan ofthe scan sequence is performed by the first receiving block. The firstscan begins at scan position 0 of the first code period (code period 0)and ends at scan position 0 of the second code period (code period 1),which is indicated by the thick solid line with an arrow spanning thecomplete time axis. When the first scan begins, the second receivingblock is idle. After a half a code period delay, indicated by the thindashed line, the second receiving block begins the second scan at scanposition 4, which is indicated by the thin solid line. Thus, at the endof the first code period of time, the first receiving block hascompleted the first scan while the second receiving block has completedhalf of the second scan.

After completing the first scan, the first receiving block delays forone quarter of the second code period and begins the third scan at scanposition 2 and completes three quarters of the scan by the end of thesecond code period. During the same code period, the second receivingblock completes the second scan at scan position 4, delays one quartercode period, and begins the fourth scan at scan position 6. The processcontinues with the two receiving blocks alternately performing theremainder of the reverse binary scans until the first receiving blockcompletes the seventh scan at scan position 3 of the sixth code period(code period 5) and the second receiving block completes the eighth scanat scan position 7 of the same code period. Collaboratively, the tworeceiving blocks complete the reverse binary scan process in a total of5.875 code periods.

Optimization of a Reverse N-ary Scanning Order For Multiple ReceivingBlocks

As was the case with a single receiving block arrangement, the order ofa reverse N-ary scanning sequence can be optimized for a multiplereceiving block arrangement in order to reduce the requirement forbackward shifting between scans and thereby reduce the time required tocomplete the scanning process. In one embodiment of the invention, thesequence of reverse N-ary scan positions is determined by setting eachscan position of a given scan resolution to an available scan positionthat is forward of the previous scan position. In a preferredembodiment, the sequence of reverse N-ary scan positions is determinedby setting each scan position of a given scan resolution to the closestavailable scan position that is forward of the previous scan position.With either embodiment, if a forward scan position is not available, theavailable backward scan position having the lowest index is selected.

The optimized reverse binary scan position ordering process for amultiple receiving block arrangement can be visualized in relation toFIG. 11K, which is exactly like FIG. 11J except with different orderingof the scan positions. In the figure, the first group of scan positionscomprising 0 and 4 are assigned to the first and second receiving blocksrespectively. The first receiving block begins the first scan at scanposition 0 and completes at the end of the first code period (codeperiod 0). During the same code period, the second receiving blockdelays for one half a code period and begins the second scan at scanposition 4. Thus, the order of the scan positions 0 and 4 in the firstgroup of scan positions is determined to be 0,4.

The order of the second group of scan positions, 2 and 6, is thendetermined. To begin the second code period (code period 1), the firstreceiving block is assigned the next available forward scan positionfrom scan position 0, which is scan position 2. Thus, after a quartercode period delay, the first receiving block begins the third scan atscan position 2. Also during the second code period, the secondreceiving block completes the second scan at scan position 4 and isassigned the next available forward scan position from scan position 4,which is scan position 6. Thus, after a quarter code period delay, thesecond receiving block begins the fourth scan at scan position 6. So,the order of the scan positions 2 and 6 in the second group of scanposition is determined to be 2,6 and the optimized reverse binarysequence has been determined through two levels of scanning resolutionto be 0, 4, 2, 6.

The order of the third group of scan positions, 1, 5, 3, and 7, is thendetermined. To begin the third code period (code period 2), the firstreceiving block is continuing with the third scan, which it completes atscan position 2. Under one embodiment, the first receiving block couldbe assigned any one of the three scan positions 3, 5, or 7, which areforward of scan position 2. While under a preferred embodiment, which isdepicted in the figure, the first receiving block is assigned theclosest forward scan position, which is scan position 3. Thus, the firstreceiving block delays for one eighth of a code period and begins thefifth scan at scan position 3. During the same code period the secondreceiving block completes the fourth scan at scan position 6. It isassigned the only available forward position of the third group of scanpositions, which is scan position 7. Thereafter, the second receivingblock delays for one eighth of a code period and then begins the sixthscan at scan position 7.

Continuing the process, the first receiving block completes the fifthscan at scan position 3 of the fourth code period (code period 3). It isassigned the only available forward scan position of the third group ofscan positions, which is scan position 5. Afterwards, it delays for onequarter of a code period and begins the seventh scan at scan position 5.During the same code period, the second receiving block completes thesixth scan at scan position 7. Because there are no longer any forwardscan positions remaining in the third group of scan positions, it isassigned the available backward scan position having the lowest index,scan position 1, which also happens to be the only remaining scanposition. Thus, the second receiving block delays for one quarter of acode period and begins the eighth scan at scan position 1 of the fifthcode period (code period 4). The first receiving block completes theseventh scan at scan position 5 of the fifth code period and the secondreceiving block completes the eighth scan at scan position 1 of thesixth code period (code period 5). Thus, according to the preferredembodiment, the optimized reverse binary sequence for two receivingblocks is 0, 4, 2, 6, 3, 7, 5, 1. The optimized sequence requires 5.125code periods to complete, which is approximately a 12% acquisition speedimprovement over two receiving blocks using the non-optimized reversebinary scan sequence 0, 4, 2, 6, 1, 5, 3, 7.

Scanning Across Received Pulse Train Boundaries

In an alternate embodiment, scanning is performed across received pulsetrain boundaries. With this approach, there are no delays between scansas depicted in FIGS. 10B and 10C. Instead, the code, which defines thetemplate pulse train, is shifted in a cyclic manner relative to the scanposition. For each scan, a last portion of the code is used to definethe template pulse train signal up to the scan position and a firstportion of the code is used to define the template pulse train signalafter the scan position, where the code is subdivided into a firstportion and a second portion based on the received pulse train boundarydefined by the scan position.

The difference between this alternative embodiment and previouslydescribed embodiments can be visualized using FIG. 10D. In FIG. 10D, thetime positions 20-1, 20-2, and 20-3 represent template pulse trainboundaries, where the code defining the template pulse train is appliedat the beginning of each template pulse train boundary. With thepreviously described embodiments of the invention, the summation resultis produced beginning at each template pulse train boundary in whichcase the time offsets 23-1, 23-2, and 23-3 correspond to delays duringwhich there are no attempts to acquire the received pulse train signal.Under this arrangement, the template pulse train boundary shifts witheach time offset and when the summation result passes the thresholdcheck, it is assumed that the given template pulse train boundarysubstantially coincides with the received pulse train boundary. Thus,each template pulse train boundary also represents a candidate receivedpulse train boundary, and when the threshold check is passed, the actualreceived pulse train boundary is assumed to have been substantiallydetermined and the template pulse train timing is assumed to besubstantially aligned.

With the alternative embodiment, time positions 20-1, 20-2, and 20-3continue to represent candidate received pulse train boundaries, but nolonger represent template pulse train boundaries except for the 0 scanposition case. Additionally, the time offsets 23-1, 23-2, and 23-3 nolonger correspond to delays during which there are no attempts toacquire the received pulse train signal. Instead, time offsets 23-1,23-2, and 23-3 correspond to portions of the code that “wraps”. Here,the template pulse train boundary never changes. Instead, the codedefining the pulse train shifts with each time offset with the end ofthe code wrapping to the front. With this approach, the portions of theshifted template pulse trains 20 to the right of the vertical dashedline that intersects the second and third template pulse trains willwrap around in front of the time positions 20-2 and 20-3, respectively.Accordingly, the template pulse train boundary is only assumed tocoincide with the first candidate received pulse train boundary, whichhas a time position equal to some initial time reference t0 _(ref). Forall other candidate received pulse train boundaries defined by some timeoffset from t0 _(ref), the pulse train boundary and the candidate pulsetrain boundary are assumed to not substantially coincide and thesummation result is assumed to correspond to the energy captured fromthe last portion of a received pulse train and the first portion of thefollowing received pulse train. Under this arrangement, when thethreshold check is passed and the actual received pulse train boundaryis assumed to have been substantially determined, the template pulsetrain must still be shifted by the known time offset such that itsboundary aligns with the received pulse train boundary.

An advantage of the alternative embodiment is improved acquisitionspeed. One potential disadvantage of the approach is that specializedpulse summation techniques may be required to account for crossing bitboundaries when the acquisition signal is modulated by data, whichincreases system complexity and may increase system costs.

FIG. 11L depicts reverse binary scanning across received pulse trainboundaries. In the figure, the scan positions defining the receivedpulse train boundaries are shown with a number on either side. Thenumber to the left of each received pulse train boundary represents thelast portion (or fraction) of the code used to define the template pulsetrain signal from the beginning of the code period to the boundary andthe number to the right of the boundary represents the first portion (orfraction) of the code used to define the template pulse train signalfrom the boundary to the end of the code period. Under this arrangement,a portion of a code corresponds to the code elements that specify pulsecharacteristics during the corresponding portion (or fraction) of thecode period defined by the received pulse train boundary. However, itshould be understood that the numbers of pulses of the template pulsetrain defined by the first and second portions of the code are notnecessarily proportional to the portion (or fraction) of the code perioddefined by the received pulse train boundary. For example, half a codeperiod does not necessarily represent half of the pulses of a templatepulse train, but instead equals half of the time corresponding to a fulltemplate pulse train.

With this embodiment of the invention, the first scan of the reversebinary scanning process begins at scan position 0 where the receivedpulse train boundary coincides with the beginning of the code period.For the second scan, the received pulse train boundary is defined byscan position 4. Thus, the second half of the code is used to define thetemplate pulse train from the beginning of the second code period (codeperiod 1) to scan position 4 and the first half of the code is used todefine the template pulse train from scan position 4 to the end of thecode period. Similarly, for the third scan, the received pulse trainboundary is defined by scan position 2. Here, the last quarter of thecode is used to define the template pulse train from the beginning ofthe third code period (code period 2) to scan position 2 and the firstthree quarters of the code is used to define the template pulse trainfrom scan position 2 to the end of the code period. The processcontinues for each of the remaining scans. Thus, each scan requires onecode period and the total scanning time is 8.0 code periods.

With this embodiment of the invention, there is no benefit to be gainedby reordering the reverse binary scanning sequence since there are nodelays between scans. It should also be understood that scanning acrossreceived pulse train boundaries may be used to speed acquisition timesof each of the other scanning approaches such as those described herein(e.g., coarse step scanning, fine step scanning, frame step scanning,random offset scanning, pseudo noise offset scanning, etc.) and mostother scanning approaches as would be understood by one skilled in theart.

As stated previously, specialized pulse summation techniques may or maynot be required depending on whether the transmitted pulse train signalis modulated or not and, if it is modulated, depending on thecharacteristics of the acquisition data signal.

In one embodiment, the transmitter sends an acquisition signal that isnot modulated by data for a period of time to aid acquisition. Thetransmitted signal is modulated only by a code. With this embodiment,specialized pulse summation techniques are not required, and a singlecorrelator 710 and single pulse summation stage 734 will suffice asdepicted in FIG. 7.

In another embodiment, the transmitter sends an acquisition signal thatis modulated by a constant data signal. In other words, at least aportion of the acquisition signal may be all 0 bits or all 1 bits. Withthis embodiment, specialized pulse summation techniques are again notrequired because although the received pulse train boundary alsorepresents a bit boundary, the bits on either side of the bit boundaryare the same.

In still another embodiment, the transmitter sends an acquisition signalthat is modulated by data that may vary. With this embodiment,specialized pulse summation techniques are required because both thedata bit before the bit boundary and after the bit boundary are unknown.There are four possible combinations for the bits before and after theboundary: 0 0, 0 1, 1 0, and 1 1. As previously described, if both bitsare the same, then it does not matter whether a bit boundary is scannedacross. However, if the bits are different, then the energy capturedcorresponding to one or the other bit must be inverted prior to beingsummed. Thus, the 0 1 and 1 0 bit combinations require either the energycaptured before the bit boundary or the energy captured after the bitboundary to be inverted.

In one embodiment, the impulse radio receiver is modified to have aninverting pulse summation stage capable of being switched on duringacquisition, which inverts either the energy before the bit boundary orthe energy after the bit boundary based on a control signal. Referringto FIG. 11M, the pulse summation stage 734, sample and hold 736, anddetector stage 738 of FIG. 7 are supplemented by an inverting pulsesummation stage 1174 and sample and hold 736 a that also outputs to thedetector stage 738. During signal acquisition, switch 1172 may beengaged. When switch 1172 is engaged, inverting pulse summation stage1174 inverts the energy captured before the bit boundary or the energycaptured after the bit boundary, as controlled by a control signalreceived from the precision timing generator 714, and integrates boththe inverted and non-inverted captured energy to produce an integratedoutput signal. The integrated output signal is input into sample andhold 736 a. The output of sample and hold 736 a is input into detectorstage 738 which performs a threshold check using the outputs of the twosample and holds 736 and 736 a, where the threshold check is passed ifeither sample and hold output meets the established threshold.

FIG. 11N and FIG. 11O depict alternative embodiments of an invertingpulse summation stage. In FIG. 11N, the inverting pulse summation stage1174 includes a switch 1176 controlled by the control signal, as mightbe output by precision timing generator 714, which directs an inputsignal to either output A or output B. Output A is received directly bysumming device 1180 while Output B is directed to an inverter 1178 thatinverts the input signal prior to it being input into summing device1180. Under this arrangement, switch 1176 can be controlled such thateither the energy prior to the bit boundary or the energy after the bitboundary is inverted.

In FIG. 11O, the inverting summation stage 1174 includes a multiplier1182 and a summing device 1180. The multiplier 1182 receives a positivecontrol signal, e.g., a positive square wave signal, or a negativecontrol signal, e.g., a negative square wave signal, as might be outputby precision timing generator 714, which is mixed with the input signal.When the control signal is positive, the input signal is not inverted.When the control signal is negative, the input signal is inverted. Thus,the control signal can be a negative square wave before the bit boundaryand a positive square wave after the bit boundary, or vice versa, tocause either the energy prior to the bit boundary or the energy afterthe bit boundary to be inverted.

Scanning Processes in Multipath

Large step scanning, and in particular, binary scanning can haveadvantages in a multipath environment. These advantages will now bedescribed with respect to FIG. 11P. The multipath advantages arise fromthe ability of a UWB system under certain circumstances to acquire oneor more of the reflections as an alternative to the direct path signal.Since the strongest reflections typically follow closely in time fromthe direct path signal, this creates a time interval following thedirect path signal where an acquisition step will very likely result indetection and acquisition. Conversely, an acquisition step outside thistime interval is much less likely to result in acquisition. It can thusbe observed that a sequence of steps just equal to this interval wouldfind the signal or a reflection in a minimum number of steps. It is notnecessary to step through the entire search space at ¼ cycle in order toacquire the signal.

Thus, if the multipath pattern for a channel is known, the search stepsize can be established accordingly. The multipath pattern may be knownin the case where a message has been previously communicated and the twoparties have not moved appreciably in the interval, or in the case wherelock is temporarily lost and must be reacquired. Packet modecommunications is a particular example where the multipath pattern canbe utilized advantageously.

If a system is designed with a fixed large step scanning process andthere is no multipath or the multipath interval is smaller than the stepsize, there exists a possible condition where the system may step overthe signal and not acquire in an acceptable time. This may be correctedby assuring that for each rescan of the signal space that someadditional offset, which may be ¼ cycle or some random pattern or otherpattern, is added to insure that the offsets within the step size areeventually sampled.

The reverse binary scanning approach has advantages of both large stepand small step scanning and is particularly adapted to the situationwhere the multipath is unknown and may vary over a wide range. In thisregard, the reverse binary scanning approach utilizes the whole searchlength as the initial step size by way of the zero sample and thendivides this length in half, then further divides the half in half, buteach succeeding half is not sampled in sequence, but itself is sampledwith a reverse binary pattern.

Thus, the reverse binary pattern quickly presents a new transmissionwith a maximal width step successively dividing in half no matter wherethe new signal begins in the cycle. Thus, for any multipath intervalwidth, the search space will be searched to that width based on thebinary halving process beginning from the start of transmission ratherthan from the start of the reverse binary sequence. For example, areverse binary sequence with an ultimate resolution of 16 bits will take65536 steps to complete a cycle. The equivalent fine resolution systemwould take 65536 steps to complete a cycle. For a multipath interval of1/16 of the search space, the reverse binary system could search theentire search interval to that resolution in 16 steps beginning with anyone of the 65536 possible starting steps. Each sample would have a 1 in16 chance of landing in the multipath interval. In contrast a fineresolution linear scan system would have a 1 in 16 chance of starting inthe multipath interval, but if not, it may take on average half of theremaining search interval to reach the multipath interval, in this casebeing ½(65536−4096)=30720 steps.

Because of the halving process, the reverse binary method matchesinherently to any multipath interval size from minimum resolution to thefull width of the scan interval without prior knowledge of the channelmultipath characteristics.

The disclosure herein teaches structures with properties similar to thereverse binary pattern and identifies key properties that will enableone skilled in the art to design similar patterns, thus it is consistentwith the intended scope of the invention to include sequences that havesimilar or equivalent structure and method to the reverse binarypattern.

The Acquisition Process (Continued)

Referring back to FIG. 9, in block 3, a correlation result is obtainedfor each pulse in the template pulse train. The received impulse radiosignal and the template pulse train are compared via a cross-correlator(e.g., multiplication and integration, or a multiplication andsummation) as illustrated with FIGS. 8A-8C.

In block 4, the output of block 3 is further summed or integrated, ifnecessary, to yield a summation result. Many types of modulation can beused to convey data using UWB signals as are known in the art ordescribed in documents incorporated herein. The present invention is notlimited to the modulation types described as examples herein, but may beadapted by one skilled in the art to other types of modulation.

FIG. 12A illustrates an exemplary output of an integrator providing asignal summation of multiple pulse signals from the output of thecorrelator or sampler of block 3. Basic binary modulation utilizingmultiple pulses per digital bit is shown in the example illustrated inFIG. 12A.

Referring to FIG. 12A, the integrator integrates signal plus noise overan integration interval comprising a single digital bit of information.In the absence of noise and with equal pulse amplitudes for the pulsesin the interval, the output of the integrator will be a stair step orramp. Shown in the figure is the output with signal plus noise showingvariation from the linear ramp shape due to noise. Shown in the figureare multiple ramp signals overlaid as might appear on an oscilloscopesynchronized with the data bit clock to show multiple traces overlaid onthe same display. Also shown are positive and negative ramps as would begenerated by a flip or time shift modulation where positive ramps may beassociated with one digital value such as “1” and the negative ramps maybe associated with another digital value such as “0”. At the end of theintegration interval, the integrator may be dumped to ready theintegrator for the next bit of information. Before dumping theintegrator output, the integrator output is tested to determine if it ispositive or negative. A positive value is registered as a “1” and anegative value is registered as a “0”.

FIG. 12B illustrates a statistical probability density relating to theoutput and final value of the summation of FIG. 12A. FIG. 12Billustrates a probability density function for the positive ramp outputvalues as a function of voltage and a probability density function forthe negative ramp output values as a function. The figure is oriented toutilize the same voltage scale as FIG. 12A to illustrate thecorrespondence between the ramp ending values and the statistical meanand standard deviations shown in FIG. 12B. Each density function has amean and a standard deviation. For signals received in thermal noise thedistribution is typically assumed to be Gaussian. For interference, thedistribution may be some other function.

In block 5, a threshold check of the comparison result, or the summationresult, from block 4 is performed. Shown in FIGS. 12A and 12B is athreshold value T which may be used to estimate whether a signal ispresent and in synchronization or not present and\or not insynchronization. Preferably, the summation result from block 4 iscompared to the threshold T. If the magnitude of the summation result isgreater than or equal to the threshold T, it is assumed the templatepulse train has matched the impulse radio signal. If the magnitude ofthe summation result is less than the threshold T, it is assumed thatthe template pulse train does not match the impulse radio signal.

In a preferred embodiment, FIGS. 12A and 12B represent a design minimumsignal strength. A nominal value for the associated threshold is shownin FIGS. 12A and 12B as ½ of the design minimum signal strength. Othermethods of setting threshold T are possible such as a factor based onnoise variance or a level established by a constant false alarm controlloop. If the design minimum signal strength is established as a meansignal-to-RMS noise of 10 dB or about 3/1, the threshold would be set atabout 1.5/1, which would result in about one false alarm in 16 bits forGaussian noise. Such a high value suggests that false alarms should beidentified within a very few data bit times. It can be appreciated thatif it takes 16 bit times to identify false alarms and false alarms areproduced at an average rate of one in 16 bit sample times, the locksearch time would be cut in half. Other thresholds can be establishedthat are more or less sensitive to noise. This example is used toillustrate that a fast method of identifying and eliminating falsealarms enables lower threshold values and thus enables fasteracquisition for lower signal levels.

In block 6, if the comparison result passed the threshold check, flowproceeds to block 7. If the comparison result failed the thresholdcheck, flow proceeds back to block 2.

In block 7, after the comparison result passes the threshold check, alock loop is engaged to track the timing of the received signal.Engaging the lock loop is also referred to as “locking on” the signal or“closing” the lock loop. Preferably, the lock loop is engaged at theearliest practical point in the flow and may be engaged as early asBlock 2 of FIG. 9. Early closing of the lock loop minimizes the openloop time during which the receiver reference oscillator may driftrelative to the transmitter of the desired signal. Excessive drift cancause the impulse radio signal to be lost, and the process to berestarted. There can be a variety of systems used to lock and track thereceived impulse radio signal. As examples, the following systems can beused for locking on the received impulse radio signal: phase lockedloop, decision feedback loop, Costas loop, and other UWB lock loopsdescribed in the art. Several such techniques are described in U.S. Pat.Nos. 5,812,081 and 5,832,035, which have been previously incorporated byreference.

1/t Gain Scheduling in a Lock Loop

FIG. 13A illustrates a system for use in locking and tracking thereceived impulse radio signal. The system in FIG. 13A employs two gainsblocks 1304 and 1308 and two delays 1312 and 1316. The closed-loopsystem employs a proportional-integral controller 1340, which includesgain 1304, gain 1308, and delay 1312. The values for the gains 1304 and1308 can be determined and fixed to obtain desired control loopproperties. As an option, the gain values for the gains 1304 and 1308can have a time varying component. This component is illustrated as gainfunctions 1320 and 1324. In a preferred embodiment, these gains may beF₁(t)=1/sqrt(t) 1320 and F₂(t)=1/t 1324.

The operation of the invention will now be described in greater detailwith reference to FIG. 13A. FIG. 13A is a simplified diagramrepresenting the signal flow in a discrete time lock loop in accordancewith the present invention.

Referring to FIG. 13A, signal E₁ and signal N₀ represent the signal andnoise components respectively of a received signal as presented to theinput of a sampler in a receiver system. The combined signal 1326 issampled by the sampler 1328 in accordance with timing controlled bysignal E_(t). E_(t) is a timing command signal which results from thecontrol loop calculation. Sampling of E₁ at time E_(t) produces acontrol error signal 1330. In a preferred embodiment, the operation ofthe control loop is at an update rate synchronous with and equal to thedata rate. Thus the delays 1312 and 1316 are equal to the data bit timein, for example, a flip modulated system. Accordingly, the samplingfunction 1328 may include the sampling and integrating of multiplepulses comprising one bit's worth of error signal and may includedecision feedback as appropriate. The system, however, may be configuredto operate at an update rate that is less than or more than the systemdata rate by appropriate adjustment of the gains.

After sampling and summing, the error signal 1330 is then optionally,but preferably, adjusted 1332 to compensate for signal amplitude asdetermined by a measurement of signal amplitude 1334. In a typical UWBreceiver, the tracking loop is configured to track a zero crossing ofthe received signal. A transfer function relating the timing commandsignal E_(t) to the sampler output error signal 1330 includes a timeerror to voltage out function that is proportional to the receivedsignal E₁ input. Thus, in a preferred embodiment, the amplitudedependency is compensated in block 1332. The resulting amplitudecompensated signal 1336 is then passed through two paths. A first gainpath scales the signal by a first constant 1304 and optionally scalesthe signal by a first time varying value 1320. A second path scales thesignal by a second constant 1308 and also optionally scales the signalby a second time varying value 1324. The second path includes a firstdelay 1312 and first feedback summation 1314.

The output of the first path 1332 and the output of the second path 1334are summed to produce a signal representative of a frequency controlsignal E_(f). This frequency control signal E_(f) is then passed througha feedback summing function comprising a second delay 1316 and summation1318, which outputs the time control signal E_(t) which commands thetiming for subsequent samples to correct for the error in accordancewith the error signal and thus completes the loop. In one alternativeembodiment of the invention, the frequency control signal E_(f) iscoupled to the frequency control input of an alternative precisiontimer, the alternative precision timer being designed to be controlledby a frequency control E_(f) rather than a time control E_(t).

Referring to FIG. 13A, two gain blocks are indicated where the gain is afunction of time F₁(t)1320 and F₂(t)1324. Time is measured from theinstant of closing of the loop. These gain blocks together with the gainblocks k1 1304, k2 1308 and the gain transfer function for the samplerand amplitude correction determine the dynamics of the closed loop. Apreferred dynamic for this loop response is a critically dampedresponse. Since the gain transfer function of the sampler is typically afunction of signal amplitude, the 1/A (where A represents signalamplitude) block 1332 is provided to compensate for this dependence onamplitude. Thus, the desired dynamics can be held relatively constantover a range of signal amplitudes. Since the tracking loop is typicallybased on a zero crossing of the signal, the amplitude of E_(t) is notdirectly available from signal 1330; however, amplitude may typically bederived from the data demodulation process.

The purpose of scheduling the gain as a function of time is toaccommodate two opposite extremes in the desired performance of thetracking loop and gracefully transition between these two extremes. Onone extreme, for the very first few samples, the response of the loopshould preferably be fast to capture lock in the presence of arelatively large frequency difference between the transmitter andreceiver. This is desirable because tolerance of larger frequencydifferences in the pulse rate allows lower cost devices. Conversely, therequirement for extremely tight tolerance on pulse rate frequency can beprohibitively expensive for some applications.

On the opposite extreme, once the initial frequency difference iscaptured, it is desirable to filter the tracking signal as much aspossible to minimize susceptibility to noise and interference. Theeffect of noise may be to cause loss of lock or to cause increased errorrate through timing jitter introduced at the E_(t) or E_(f) controls.The need for reduced noise suggests a preference for a slowly respondingcontrol loop. The functions F₁(t) and F₂(t) provide for these twoextremes and provide a transition from one to the other. A preferredtransition function for F₁(t) is 1/sqrt(t) where t begins at thebeginning of the first sample following the closing of the control loopso that t is equal to one sample time (the time between at the end ofthe first sample where the value of t is first utilized, and t is equalto two sample times at the end of the second sample and so on. Apreferred transition function for F₂(t) is 1/t for this particularsecond order control loop because when the two gain factors are kept inthis ratio, the dynamics of the control loop remain critically dampedover the full range of gain transition. The initial value of the gainfunction may be predetermined for a period or may be adjusted bysubstituting t=t+a, where a is an initial value. One skilled in the artwill appreciate that other similar methods may be utilized to optimizethe initial performance.

It is also preferred that the transition functions 1320 and 1324 belimited to some predetermined minimum value for large values of t.Factors influencing the choice of the minimum value include: short termstability of the transmitter and receiver reference oscillators, Dopplermotion between the transmitter and receiver, and the stability of amultipath environment that may be present. Methods for limiting thevalue of F₁(t) include simply limiting the value, and limiting t, andalternate equations for F₁(t) such as F₁(t)=b+1/t, and other similarmethods available to one skilled in the art.

FIGS. 13B and 13C represent continuous system embodiments of theinvention. FIG. 13B is similar to FIG. 13A except that the delay 13121316 and summation 1318 1314 blocks are replaced with integration blocks1342 and 1344. The operation is otherwise similar to that of FIG. 13A.

FIG. 13C is configured with the time dependent gain functions 1320 1324in series. The series configuration may allow implementationefficiencies in that the two functions may be identical allowing thesame function block 1320 1324 to be called by both calculations or thecopying of the components for one block to implement the other.

Quick Check

In block 8, after the received impulse radio signal has acquired lock, aquick check of the received impulse radio signal is performed. A quickcheck is used to quickly distinguish a false detection due to noise froma true signal lock. The quick check tests a number of subsequentportions of the impulse radio signal to verify the acquisition of thereceived impulse radio signal. The quick check preferably removes anyfalse alarms signaled in block 6. The quick check determines that atleast X of the next Y tested portions of the impulse radio signal matchthe template pulse train. For example, X=2, and Y=3.

FIG. 14 illustrates a flow diagram for the quick check of block 8. Thequick check incorporates several of the blocks from FIG. 9 anddetermines whether the template pulse train is aligned with the receivedimpulse radio signal via a repetition of blocks 2-6 in FIG. 9. In block45, the quick check starts. In block 46, a template pulse train counteris set to 0, and a pass counter is set to 0. In block 47, the templatepulse train counter is incremented by 1. Blocks 49-52 are the same asblocks 3-6 in FIG. 9, and the discussion of these blocks is omitted.

In block 53, the pass counter is incremented by 1. In block 54, thevalue for the pass counter is compared to X, where 1≦X. If the passcounter is equal to X, flow proceeds to block 55. If the pass counter isnot equal to X, flow proceeds to block 56. In block 55, the receivedimpulse radio signal passed the quick check.

In block 56, the template pulse train counter is compared to Y, where1≦X≦Y. If the template pulse train counter is equal to Y, flow proceedsto block 57. If the template pulse train counter is not equal to Y, flowproceeds to block 47. In block 57, the received impulse radio signalfailed the quick check. In block 58, the quick check is finished.

Data Synchronization

Referring to FIG. 9, after the quick check is performed, flow proceedsto block 9. In block 9, it is determined if the received impulse radiosignal passed the quick check. If the received impulse radio signalpassed the quick check, flow proceeds to block 10. If the receivedimpulse radio signal failed the quick check, flow proceeds to block 2.

In block 10, a synchronization check of the received impulse radiosignal is performed. Once the quick check has been passed, it is assumedthat the locations of the pulses in the received impulse radio signalhave been determined. With the synchronization check, the location ofthe beginning of the acquisition data is determined.

FIG. 15A illustrates acquisition data 25. The acquisition data 25 istransmitted by the transmitter and includes synchronization data 26 andcommand data 27. Preferably, the synchronization data 26 is determinedby an error tolerant code, such as one of the Stiffler codes. Stifflercodes are described in the following, which is incorporated herein byreference: J. J. Stiffler, “Synchronization Techniques,” in DigitalCommunications with Space Applications, edited by S. W. Golomb, 1964,pp. 135-160. For example, if four bits are used for the synchronizationdata, one of 16 code words, as determined by the Stiffler code, can beused for the synchronization data 26. Advantageously, by using Stifflercodes, one code word is not mistaken for another code word. Preferably,the synchronization data 26 has approximately 32 data bits, and thecommand data 27 has a length of approximately 32 data bits. The commanddata 27 is discussed further with respect to FIG. 18. In addition, withthe invention, Stiffler codes are used for encoding an impulse radiosignal to lock and acquire the impulse radio signal.

FIG. 15B illustrates a flow diagram for the synchronization check ofblock 10. In block 61, the synchronization check starts. In block 62, asynchronization counter is set to Z, where Z is a natural number and1≦Z. In block 63, Z bits of the received impulse radio signal areacquired. In block 64, it is determined whether the Z bits match astored synchronization code. If the bits match, flow proceeds to block65. If the bits do not match, flow proceeds to block 66. In block 65,the signal passed the synchronization check.

In block 66, it is determined whether the synchronization counter isequal to V, where V is a natural number and 1≦Z≦V. If thesynchronization counter is equal to V, flow proceeds to block 69. If thesynchronization counter is not equal to V, flow proceeds to block 67. Inblock 67, the synchronization counter is incremented by 1. In block 68,an additional bit is acquired, and flow proceeds to block 64. In block69, the signal failed the synchronization check. In block 70, thesynchronization check is finished.

If the received impulse radio signal passed the synchronization check,the beginning of the acquisition data has been located, and if thesignal failed the synchronization check, the beginning of theacquisition data has not been located. By selecting V for block 66, thenumber of bits checked prior to failure is determined.

Referring to FIG. 9, the flow proceeds from block 10 to block 11. Inblock 11, it is determined whether the received impulse radio signalpassed the synchronization check. If the received impulse radio signalpassed, flow proceeds to block 12, and if the received impulse radiosignal failed, flow proceeds to block 2.

Command Check

In block 12, a command check of the received impulse radio signal isperformed. In the command check, it is assumed that the beginning of theacquisition data 25 transmitted by the transmitter has been determinedvia the synchronization check. In block 12, it is determined whether thecorrect acquisition data has been received by the receiver and whetherthe receiver has acquired the correct impulse radio signal.

Referring to FIG. 15A, the contents of the command data 27 of theacquisition data 25 determines the outcome of the command check. Forinstance, the command data 27 of the acquisition data 25 can include atransmitter identification and/or a receiver identification. Thereceiver may store a transmitter identification corresponding to thetransmitter from which information is being received, and may store areceiver identification identifying itself. If the transmitteridentification of the command data 27 does not match the storedtransmitter identification, the received impulse radio signal failed thecommand check, and otherwise, the received impulse radio signal passedthe command check. Further, if the receiver identification of thecommand data 27 does not match the stored receiver identification, thereceived impulse radio signal failed the command check, and otherwise,the received impulse radio signal passed the command check. The commanddata 27 may include the following, which may also be stored in thereceiver: transmitter identification; receiver identification; dateidentification; time identification; communication protocol; networkprotocol; power control information; and any combination thereof.

FIG. 16 illustrates a flow diagram for the command check of block 12. Inblock 75, the command check starts. In block 76, the command data of thereceived impulse radio signal is determined. In block 77, the commanddata from the received impulse radio signal is compared to storedcommand data. If the received command data matches the stored commanddata, flow proceeds to block 78. If the received command data does notmatch the stored command data, flow proceeds to block 79. In block 78,the signal passed the command check. In block 79, the signal failed thecommand check. In block 80, the command check finishes. Referring toFIG. 9, flow proceeds from block 12 to block 13. In block 13, it isdetermined whether the received impulse radio signal passed the commandcheck. If the received impulse radio signal passed, flow proceeds toblock 14, and if the received impulse radio signal failed, flow proceedsto block 2. In block 14, the received impulse radio signal has beenacquired and locked on. In block 15, the fast lock and acquisition ofthe impulse radio signal is finished.

To assist the receiver in maintaining lock and to reduce the necessityof a re-lock and re-acquisition period after the impulse radio signal islost by the receiver, locking and acquisition data can be interspersedamong the information data sent by the transmitter. For example,additional acquisition data 25 can be interspersed by the transmitteramong the information data. Depending on the type of locking andacquisition data interspersed, same or all of the aspects of the processillustrated in FIG. 9 are performed when the locking and acquisitiondata is transmitted by the transmitter.

Radio System

FIG. 17 illustrates a block diagram for a system configured inaccordance with the present invention. The system of FIG. 17 can beimplemented in a receiver for a one-way or two-way communication system.

The system of FIG. 17 is similar to the system of FIG. 7, except forblocks 1704, 1707, 1708, 1709, and 1711 that are used to implement fastlock and acquisition of the impulse radio signal. During the searchphase of acquisition, the time base is initially controlled directly bythe controller through switch 1711. After the time position and rate areestablished, the lock loop may optionally be engaged using switch 1711to connect the output of the lock loop filter to the input of the timebase. The engaging of the lock loop may alternatively occur at a latertime such as, for example, after threshold detection or after quickcheck; however, later engaging of the lock loop typically requires moreaccurate time base oscillators. Referring to FIG. 17, the controller1709 initially commands switch 1711 through the “C” command input toconnect switch output “O” to input “B” thereby allowing the controller1709 to control the time base 718 directly. Once the time base 718 timeposition and frequency are set, the controller 1709 may command theswitch 1711 to connect switch input “A” to output “O” in accordance withthe threshold compare 1704 and detector 1706 signals to “close” the lockloop and track the received signal.

A discussion of the components of FIG. 17 that are identical to those ofFIG. 7 is omitted for clarity.

In an alternative embodiment, two correlators may be used to provideseparate signals for tracking and data. FIG. 18 illustrates a receiverutilizing two correlators to provide separate signals for tracking anddata. Referring to FIG. 18, the received signal from the antenna iscoupled to a data path and a tracking path. The data path comprisingcorrelator 704 and additional blocks coupled thereto. The tracking pathcomprising correlator 1802, the lock loop filter 1705 and other blockscoupled thereto. The two correlators are fed template 728 signalsseparated in time by a delay 1804, which is typically about ¼ cycle of asine wave at the center frequency of the UWB signal. The tracking pathis configured to track on a zero crossing of the received signal therebyplacing the data correlator in proper time position to receive a maximumon the received signal.

The template generator 728, the precision timing generator 714, the codesource 722, and the adjustable time base 718 implement a template pulsetrain generator 1710. Other techniques for implementing the templatepulse train generator 1710 can be used within the scope of theinvention.

The cross-correlator 710 performs the function in block 3 of FIG. 9. Thecross-correlator 710 combines a received impulse radio signal from theantenna 704 and a template pulse train from the template pulse traingenerator 1710.

The pulse summer 734 performs the function in block 4 of FIG. 9. Ingeneral, the pulse summer 734 produces an increasing ramp function 30 ordecreasing ramp function 31 as in FIG. 12A when the template pulse trainprovided by the template pulse train generator 1710 aligns with thereceived impulse radio signal and otherwise produces a signal that isapproximately equal to the summation of positive and negative randomnumbers or equal to a noise-like signal.

A threshold comparator 94 receives the output of the pulse summer 734and performs the function of block 6 in FIG. 9. The output of thethreshold comparator 1704 is provided to a controller 1709.

A lock loop filter 1705 receives the output from the cross-correlator710. The lock loop filter performs the same function as the lock loopfilter 742 in FIG. 7. In addition, the lock loop filter 1705 performsthe function of block 7 in FIG. 9 and locks on the received impulseradio signal. The output of the lock loop filter 1705 is provided to thecontroller 1709.

A detector 1706 receives the output from the pulse summer 734. Thedetector 1706 performs the same function as detector 738 in FIG. 7. Inaddition, the detector 1706 is coupled to a synchronization data memory1707 and a command data memory 1708. The detector 1706 is used toperform the synchronization check in block 10 using the synchronizationdata memory 1707 and the command check in block 12 using the commanddata memory 1708.

The controller 1709 receives the outputs from the threshold comparator1704, the lock loop filter 1705, and the detector 1706. The controller1709 performs the quick check in block 8. In addition, the controlleroversees the operation of the threshold comparator 1704, the lock loopfilter 1705, and the detector 1706.

As an option, the synchronization data memory 1707 can be coupled to thecontroller 1709, instead of to the detector 1706. With this option, thecontroller 1709 performs the synchronization check in block 10 using thesynchronization data memory 1707.

As an option, the command data memory 1708 can be coupled to thecontroller 1709, instead of to the detector 1706. With this option, thecontroller 1709 performs the command check in block 12 using the commanddata memory 1708.

It should be understood that the system can be configured to includeadditional sets of blocks 710, 734, 1704, 1706 and 1709 that wouldenable multiple functions of blocks 2-6 of FIG. 9 to be performed inparallel. For instance, if there were four sets of blocks 710, 734,1704, 1706 and 1709, then during the same time period four differentshifted template pulse trains can be compared to the received impulseradio signal.

The system for FIG. 17 can be implemented with, for example, thefollowing: circuitry; software and a microprocessor, microcontroller, orsimilar device or devices; and any combination thereof.

The invention has been described as using counters, for example in FIGS.14 and 15B. As those skilled in the art will recognize, other techniquescan be used for counting in addition to those specifically discussedhere, and these other techniques are included within the scope of theinvention.

The invention has been described as using a single correlator 710. As anoption, the invention can be used with multiple correlators, vectormodulation, flip modulation, and/or flip with shift modulation, such ashas been previously described. With multiple correlators, one or morecorrelators can be used to detect data, and one or more correlators canbe used to lock and acquire a received impulse signal, therebydecreasing the lock and acquisition period. With various modulationtechniques, such as vector modulation, flip modulation, and/or flip withshift modulation, additional data states can be used, thereby increasingdata speed.

The invention has been described in detail with respect to preferredembodiments, and it will now be apparent from the foregoing to thoseskilled in the art that changes and modifications may be made withoutdeparting from the invention in its broader aspects, and the invention,therefore, as defined in the claims is intended to cover all suchchanges and modifications as fall within the true spirit of theinvention.

1. A method for detecting an ultra wideband signal, comprising the stepsof: a) receiving an ultra wideband signal; b) establishing, using aplurality of processing blocks, a plurality of time offsets between atransmitter clock and a receiver clock, said plurality of processingblocks controlling at least one scanning process, said plurality of timeoffsets corresponding to said at least one scanning process; c)comparing, using said plurality of processing blocks, a plurality ofvalues of a parameter of the received ultra wideband signal to anacceptance criteria, each one of said plurality of values being inaccordance with a corresponding one of said plurality of time offsets;and d) if each one of said plurality of values of said parameter failsto meet the acceptance criteria, returning to step (a), and if any oneof said plurality of values of said parameter meets the acceptancecriteria, locking on the received ultra wideband signal.
 2. The methodof claim 1, step b) further comprising the step of: i) collaboratingusing at least two of said plurality of processing blocks to perform ascanning process.
 3. The method of claim 1, wherein said plurality ofprocessing blocks correspond to a plurality of receiving blocks.
 4. Themethod of claim 1, wherein said plurality of processing blockscorrespond to a plurality of cross-correlators.
 5. The method of claim1, wherein said at least one scanning process comprises at least one ofa fine step scanning process, a coarse step scanning process, a reverseN-ary scanning process, a frame step scanning process, a random offsetscanning process, a pseudo noise offset scanning process, or a mixedbinary scanning process.
 6. The method of claim 1, wherein saidacceptance criteria comprises one of meeting an acceptance threshold,passing a quick check, passing an synchronization check, or passing acommand check.
 7. The method of claim 1, step b) further comprising thestep of: i) varying said at least one scanning process using one of anexclusive or-ing operation, a ones complement operation, a twoscomplement operation, and a sequence order reversing operation.
 8. Areceiver for detecting an ultra wideband signal, comprising: at leastone receiving block that receives an ultra wideband signal; and aplurality of processing blocks that establish a plurality of timeoffsets between a first clock of a transmitter of said ultra widebandsignal and a second clock of said receiver, said plurality of processingblocks controlling at least one scanning process, said plurality of timeoffsets corresponding to said at least one scanning process, saidplurality of processing blocks comparing a plurality of values of aparameter of the received ultra wideband signal to an acceptancecriteria, each one of said plurality of values being in accordance witha corresponding one of said plurality of time offsets, said receiverlocking on the received ultra wideband signal when any one of saidplurality of values of said parameter meets the acceptance criteria. 9.The receiver of claim 8, wherein at least two of said plurality ofprocessing blocks collaborate to perform a scanning process.
 10. Thereceiver of claim 8, wherein said at least one receiving blockcorresponds to a plurality of cross-correlators.
 11. The receiver ofclaim 8, wherein said at least one scanning process comprises at leastone of a fine step scanning process, a coarse step scanning process, areverse N-ary scanning process, a frame step scanning process, a randomoffset scanning process, a pseudo noise offset scanning process, or amixed binary scanning process.
 12. The receiver of claim 8, wherein saidacceptance criteria comprises one of meeting an acceptance threshold,passing a quick check, passing an synchronization check, or passing acommand check.
 13. The receiver of claim 8, wherein said at least onescanning process is varied using one of an exclusive or-ing operation, aones complement operation, a twos complement operation, and a sequenceorder reversing operation.
 14. The method of claim 13, step b) furthercomprising the step of: i) varying said at least one scanning processusing one of an exclusive or-ing operation, a ones complement operation,a twos complement operation, and a sequence order reversing operation.15. A method for detecting an ultra wideband signal, comprising thesteps of: a) receiving, using a plurality of receiving blocks, an ultrawideband signal; b) establishing, using at least one processor, aplurality of time offsets between a transmitter clock and a receiverclock, said plurality of time offsets corresponding to at least onescanning process; c) comparing, in parallel, a plurality of values of aparameter of the received ultra wideband signal to an acceptancecriteria, each one of said plurality of values being in accordance witha corresponding one of said plurality of time offsets; and d) if eachone of said plurality of values of said parameter fails to meet theacceptance criteria, returning to step (a), and if any one of saidplurality of values of said parameter meets the acceptance criteria,locking on the received ultra wideband signal.
 16. The method of claim15, step b) further comprising the step of: i) collaborating using atleast two of said plurality of receiving blocks to perform a scanningprocess.
 17. The method of claim 15, wherein said plurality of receivingblocks corresponds to a plurality of processing blocks performed by saidat least one processor.
 18. The method of claim 15, wherein saidplurality of receiving blocks correspond to a plurality ofcross-correlators.
 19. The method of claim 15, wherein said at least onescanning process comprises at least one of a fine step scanning process,a coarse step scanning process, a reverse N-ary scanning process, aframe step scanning process, a random offset scanning process, a pseudonoise offset scanning process, or a mixed binary scanning process. 20.The method of claim 15, wherein said acceptance criteria comprises oneof meeting an acceptance threshold, passing a quick check, passing ansynchronization check, or passing a command check.